BACKGROUND: There is always discussion on how to ground a shield. The answer lies in what the
worst-case noise situation is. It isn’t always the same answer. Do you leave one end open or
ends? If you ground only one ends, which end?
We can look at how the interference behaves to answer these questions and they follow understood electrical properties within swept frequency regions.
BODY: The first step is to understand what a shield is doing, and how. In the simplest terms, a shield creates two separate “electrical” environments, one on each side of the shield. One side is measured as a RATIO of the field’s intensity relative to the other side, in dB.
The shields we are working with are ONLY effective with an EM wave that is predominantly “E” field in nature versus “B” field, or magnetic. Magnetic fields are not shielded or blocked with conductive shields, but need a shield material that blocks magnetic flux lines or a low permeability material…think “material a magnet will stick to.” Those kinds of material allow magnetic flux lines an easier path than through air. We can capture and re-route the flux lines in low permeability materials. For this discussion, we will look at mostly electric field shields, stuff that conducts electricity.
What is an ideal electrical shield? It is a shield that 100% blocks electrical energy at the surface of the shield,
that has infinitely low resistance. Shields don’t have infinitely low resistance and they block electrical
energy at differing impedance based on the skin depth of the shield at a specific frequency.
We can measure the effectiveness of a shield across frequency with a transfer impedance plot. This is measured in milli-ohm/meter. It describes the resistance we can expect a shield to have, and thus the ratio of the fields energy in the shield based on the CURRENT the resistance causes to flow at that frequency…and it is not linear.
The graph below shows the frequency-dependent nature of a shield. A perfect shield would have NO RESISTANCE and both ends would be identical and thus seem like a SINGLE point of reference to a flow of current. Since we have zero resistance across the shield, we can’t have current flow caused by the shields. We CAN have current flow between the two points connected at the ends of the shield. In a “perfect” world the GROUND at both ends is the same potential and thus, an ideal shield has ZERO current flow, and is a measure of the POTENTIAL on one side of the shield relative to the other, in dB.
Since we don’t have a perfect world, Transfer Impedance describes what to expect at frequencies based on the shield’s impedance, and how that shield resistance creates a CURRENT flow and thus a voltage (shield resistance times the shield’s impedance = a voltage). When we have different resistances at each end of a shield we have current flow.
Bonded Foil +60% braid
Quad Shield 60% +40% Braid
Bonded Foil +95% Braid
The chart below is what JUST the shield impedance looks like for a set length of cable at lower frequencies. We see the same non-linear behavior of shield and frequencies.
A SEED (Shield Effectiveness Evaluation Device per IEC 61196-1) test shows the dB relationship to a Lower Shield Impedance. Series number 5 with a 95% coverage 45-degree braid and Duofoil tape is clearly superior.
Okay, we can see a shield is not perfect, and not linear. So how does this say what to do with each end of a shield? We have to weigh the CHOICE of HOW the shield WORKS to decide our fate.
– If you have ideal grounds and meet IEEE bus bar grounding (see the 568C.2 or later grounding specifications) limits it means BOTH ends can be grounded and the shield current will inductively couple less interference than the shield ATTENUATES through its material composition.
– If we have severe ground differential, we can induce a strong current in the shield that CAN, if the shield’s resistance is higher, induce noise into the core that is WORSE than if we disconnect one end. We convert our shield to an antenna, not a shield!
– An antenna does NOT create two separate environments between them with the ratio of one
measured to the other. One end of the antenna is infinite impedance (the open end) with the other end at ground.
The antenna “wire” is as close to zero impedance as possible in order to NOT
attenuate the antenna’s signal going to ground. The signal won’t go to the open end, but seeks the lowest potential in the circuit.
We trade the noise caused by a poor ground resistance potential between shield ends for the induced noise in an antenna’s wire parallel to the signal wires that induces a voltage based on the antenna resistance. In an antenna type ground, it is best to ground the SEND end, as the SIGNAL on the internal wires is as LARGE as possible relative to the antenna signal, improving the signal to noise between the two.
An often-ignored aspect of shields is HOW to ground one at lower frequency versus RF. There is a big difference and again, it is based on the shield characteristics at each frequency.
The charts below are derived from TWO slightly differing MODELS of RF shield inductive reactance resistance. I have this paper for those interested. But, the data is the same message in that as frequencies increase, the shield reactance goes UP. This necessitates a FULL 360 ground at the shield termination point in RF circuits. This is why good RF connectors are fully capturing the shield all the way around the cable. On your RF digital cables, use 360 degree grounds for the best true shielding.
SUMMARY – Most
systems will have proper GROUND differentials between them and thus have near
ZERO shield current. The shield relative to the signal wires will ATTENUATE outside interference. When you have poor grounds,
it may be beneficial to unhook the “receiver” end of the shield and hope
that the induced antenna current voltage is less severe than the induced voltage caused by differential
shield ground potentials. This should be a SECOND choice, not the first. A properly working shield, by
design, has a KNOWN shield dB rating that can be trusted in a proper electrical circuit.
An antenna ground’s induced voltage onto the cable is not fully described and is dependent on the GROUND proximity point and shield’s distance from the signal wires. In severe situations, it may be the best choice to mitigate noise to the lowest possible reference value as it is pretty hard to REMOVE a shield already on a cable. Some, such as coaxial cables, can’t be mitigated and need to be properly designed to EXCEED the ground differential by several orders of magnitude so as not to aggravate any ground differential.
ICONOCLAST will use double ground interconnect shields and proper DCR RCA grounds.
Power cables should also use grounds
at BOTH ends if you have a proper GROUND plane resistance such that ZERO current flows and thus you have ZERO induced voltage from differential current. An
antenna type ground CREATES a differential in each end of the “antenna” by design (one end is ideally infinity the other is ideally zero) and is thus a second choice if you have known ground issues.
One last note, those heavy 10 AWG power or more cables, may provide benefit as they induce less ground differential resistance than smaller power cords as the ground wires are larger. The circuit may not need the power delivery of a larger cable, but the lower ground resistance values may be of benefit on longer runs in marginal power grid situations. This will improve a shield’s current to nearer zero across frequencies. The dB isolation numbers values are for a proper shielded system with IEEE and TIA compliant shield differentials.
The Transfer Impedance numbers are between two-reference point probes on a shield, and DO NOT need the ground potential differences for characterizations.
(This article consists primarily of a technical piece by Galen Gareis, supplemented by some paragraphs by Kurt Denke, as noted herein by the initials “KD”)
BACKGROUND: The development of the ultra-low R, L and C ICONOCLAST™ home studio XLR cables exceeded design expectations and performance numbers reached. The patented use of true air tube dielectrics enabled electrical performance relative to size that is unattainable any other way.
The drawback to the pure performance ICONOCLAST™ design is that the ultimate electrical performance does require a mechanical limitation in the cable’s use that is wholly managed in the home audio space as performance is needed above all else, and in a more controlled and fixed installation. The move to a STUDIO and professional market seems nearly impossible to reach the same size and electricals as the home market design. This paper covers the unique and high performance electrical XLR design that also meets, and exceeds electrical expectations for such a strong, flexible, crush and impact tolerant design.
(KD: While this paper does not address the BAV RCA cable directly, I will append a note at the end to tie those in.) BODY: The most difficult task was the design move from foamed Teflon with solid Teflon belting in the original design to a MUCH more ruggedized version for the studio market. The careful design of the AIR TUBE dielectric used for home use cables improved the cable’s air tube physical performance (resisting deformation and physical changes under use and to advised bend limits). Changing the process variables (bonding the outer tube to the inner filler) and materials (use a modified POE, Poly Olefin Elastomer) allowed a remarkably tough, flexible and crush resistant core structure. The POE’s electrical characteristics were mitigated, as was the Teflon’s in the home ICONOCLAST XLR), by putting AIR between the conductor and the solid dielectric materials. Where the dielectric materials were required to touch the wire the lowest dielectric material was kept in place, Teflon) The selection of materials was important on two fronts: modulus of elasticity and dielectric properties. If either was deficient, the design would not work. ELASTIC MODULUS – The modulus of elasticity (also known as the elastic modulus, the tensile modulus, or Young’s modulus) is a number that measures an object or substance’s resistance to being deformed elastically (i.e., non-permanently) when a force is applied to it. The material has to stretch or compress but with a high force (lower modulus) applied. This is called elastic deformation. The chart below shows the graphical properties of plastic deformation. We need to work in the LINEAR region so the part will recover back to its original shape. https://en.wikipedia.org/wiki/Deformation_(engineering)#Elastic_deformation – definition. Linear elastic deformation is governed by Hooke’s law, which states: σ = xE Where σ is the applied stress, E is a material constant called Young’s modulus or elastic modulus, and x is the resulting strain. (This relationship only applies in the elastic range and indicates that the slope of the stress vs. strain curve can be used to find Young’s E). Engineers often use this calculation in tensile tests. The elastic range ends when the material reaches its yield strength. At this point plastic deformation begins.
In order to resist plastic deformation under tensile and compressive load simultaneously, a material with a high elastic modulus would be best. The material selected was DOW ENGAGE 8450:
The new part had to meet the same SIZE requirement for XLR compatibility as the TEFLON part, so material properties are extremely important. The PHOTOs below show the TEFLON part on the left and the DOW8450 part on the right. A very important design change other than the material was to BOND the outer POE material tube to the inner POE filler. This required a SOLID material to also enhance the YOUNG’s modulus of the overall part. Bonding the two parts, the filler and tube, is much more critical than it seems for the intended application of this part. In order to meet crush and impact resistance with the part, it had to be bonded to SIMULTANEOUSLY be crushed and stretched concurrently. When a tube is crushed, the sides will EXPAND in dimension. The inner filler X-design provides a high degree of tensile resistance to this elongation change, improving the part’s deformation resistance. Once the force is removed, the part pulls back into its original shape. This is why a high 1060 PSI material is needed with a LARGE elongation percentage: 750%.
The pictures below illustrate the TUBE’S properties under compression. With no filler in the center, the tube will crush as easily as the material properties allow. To SIGNIFICANTY improve the crush values, an inner BONDED member is added across the lines of force.
The inner member resists the expansion of the part due to compressive loads. The above “POE PART” adds TWO cross members for the air channels and strands the cable with a helix lay so under compression there is always a cross member restricting the parts expansion under load.
In the above example, the cross member goes under TENSION when the tube is compressed. This HOLDS the tube’s deformation in check, and more importantly, it has a MEMORY to what it was prior to crush, and the high elastic modulus pulls the part back into the original shape under normal compressive loads. The cable “heals” itself back to its original form. The core is tightly stranded to make sure there are always a cross member perpendicular to the load vector direction and / or TWO members largely perpendicular to the load vector.
Part is tightly STRANDED to place X-web in ideal location for compression performance and recovery.
When a part is tightly bent, it is under COMPRESSION (inner surface) and TENSION (outer surface). This hybrid part is ideally made to resist this type of plastic deformation, and recovery.
This hybrid part is ideally made to resist this type of plastic deformation, and recovery.
Since we had to use a SOLID material for ultimate Tensile / Elongation properties of the part the dielectric properties are going to be impacted to some degree. Is the erosion in electrical properties going to allow the overall high electrical standards that are also required of the finished part? Material samples were taken, along with the TEFLON reference and measured for dielectric properties. The tables below show the excellent measured electricals relative to TEFLON. We should see minimal electrical impacts in the Audio frequency range with the exception of CAPACITANCE which is directly tied to the dielectric material. But, the use of AIR TUBES mitigates the changes of capacitance.
The capacitive effect is logarithmic so the farther the material is from the reference metallic member the lower the capacitive effects. Twice the distance is one-fourth the effect. We have AIR inside the part, which is the best dielectric there is. Where material does contact the wire, we use a TEFLON thread to mitigate the capacitive effect of the wire, to thread, to outer POE material tube.
Where the magnetic fields are strongest, near the wire surface, we need AIR and/or minimal dielectric influences (use TEFLON thread). The outer influences decrease significantly with distance. If a surface is infinitely away from the conductors, the electromagnetic effects are zero. The use of AIR as a dielectric (constant = 1) allows “infinity” to be reached much sooner than if higher dielectric constant materials are used between ground plane (conductor to inner braid surface). The task is how to do this and allow extreme durability and flex, while meeting superb measured electricals. The careful positioning of material is necessary.
The overall GROUP delay is the superposition of ALL the material between the conductor and outer braid. The use of AIR keeps the VP and capacitance changes in control per measurements below on a braided core.
Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture: Cap @ 1 kHz Spec est: 8.0 +/- 2.0 pF/ft nom. 60485Y – 8.95 pF/ft
Inductance @ 1 kHz per ELP 424, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture:
Inductance Spec: N/A (uH/ft) 60485Y – 0.15 uH/ft
Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed.
VOP Spec est: 87 +/- 1 % 60485Y – 84.7%
JACKETED TEST of PD2882
Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
Cap @ 1 kHz Spec: 10.5 pF/ft max PDC2882 – 10.1026 pF/ft Inductance @ 1 kHz per ELP 424, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed.
VOP Spec: N/A (%) PDC2882 – 83.4%
The above tested numbers are excellent compared to the CONTROL all FEP design. The estimate capacitance is negligible with nearly the same group VP:
Super Flex Mic Cable
8.95-10.5 pF/ft (pin 2-3)
83.4-85.4% nom VP
A cable can be crush resistant and impact resistance but STILL be too inflexible for the intended use. This is called the BENDING MOMENT. Or, the amount of force to bend the cable ninety degrees around a specified reference mandrel when comparing cable types. This cable is EXTREMELY flexible and bends ninety degrees under its own weight (illustration).
SUMMARY: The PDC2882 initial design trial has been extraordinarily successful at meeting electrical and physical design parameters. The design actually well exceeds expectations using the DOW ENGAGE 8450 material properties.
The PDC2841 LLDPE design met similar electricals, but fell far short on physicals.
We can run a full extrude jacket with paper tape separator to improve cosmetics (perfectly round) and enhance crush even more.
CONCLUSION: The DOW 8450 material design is the best electrical and physical XLR cable design for heavy duty studio application with extremely high audio standard properties for sound quality. The DOW 8450 inner core member should provide industry leading sound quality with the best durability possible. The cable’s self-healing tensile modulus properties insure a higher degree of user abuse than any air core cable made today.
Best dielectric near the wire is NO dielectric.
We use AIR tubes, and tubes are usually STIFF by design. o 25 AWG signal wire insure high flexibility with a tight stranding lay.
A special physical and electrical material was sought after and found for this design.
Our BONDED perimeter air TUBE core solves the CRUSH and recovery issue.
Air tubes are now an ADVANTAGE in a flex application.
The target electricals were to be as near the all Teflon ICONOCLAST version as possible.
We met that goal and with FAR, FAR superior physicals; o Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture o Cap @ 1 kHz Spec: 10.5 pF/ft max PDC2882 – 10.1026 pF/ft o Inductance @ 1 kHz per ELP 42C, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture o Inductance Spec: NA (uH/ft) PDC2882 – 0.1542 uH/ft o Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed. o VOP Spec: NA (%) PDC2882 – 83.4%
Fits Standard XLR with a 0.350” outer dimension.
Cable exhibits standard XLR benefits of reduced external noise with passive CMRR design.
Air dielectric insures the best possible conductor interface.
Smaller signal wire insure better signal coherence (more even current distribution with respect to frequency)
Star quad design double-up the CMA are to act like a larger wire for longer distances.
CMRR keeps ingress noise low, even magnetic origin, with a tight strand lay (even exposure to external noise).
90% BC braid lowers EMI / RFI noise BEFORE it encounters CMRR on the four signal wires. o Wire should be purely resistive for time coherence at audio but it isn’t; – Low capacitance keeps reactive first order filter roll-off values well away from the audio band. – Low inductance improves reactive current related variables to a minimal reactance.
The paper as written, above, by Galen Gareis does not directly address the RCA cable design, but the rationale is very much the same. As Galen sets out in his Iconoclast RCA and XLR design briefs, the electricals for the RCA really ought to track those for the XLR, and in the case of the Iconoclast product they do — the RCA design is very much like a quarter-section of the XLR, eliminating two of the four quarters because the cable is unbalanced rather than balanced, and the last because the purpose of paired conductors on each polarity in a star quad is lost when common mode noise rejection is lost, as in the transition from balanced to unbalanced. This same design principle has been followed in the BAV product. The BAV RCA has a single air tube and filament, like one chamber of the BAV XLR. The resulting characteristics track those of the XLR cable as closely as possible, avoiding any characteristic difference between the sonic qualities of the two cables. )
Speaker cables are a very
different animal than high input impedance interconnecting cables. A speaker
cable connects to an extremely inconsistent 2-32 ohm (or even lower and
higher!) reactive load created by the speaker. RCA and XLR interconnect cables
see a much more consistent and resistive high impedance load making their
electrical measurements far easier to predict. The speaker cable also suffers
from the audio band’s velocity of propagation non-linearities seen in the
interconnect, but also has to figure out a way to be LOWER in impedance to
better match the speaker load, while the velocity of propagation is going DOWN,
and this naturally increases the cable’s impedance. How is all this managed as
best we can? This paper is a walk
through on how ICONOCLAST™ speaker cable addresses some of these issues.
SOUND DESIGNS CREATE SOUND PERFORMANCE™
SPEAKER CABLE DESIGN BRIEF
Shield material and design considerations.
Jacket design and material considerations.
For speaker cables, the first issue that has to be
decided is how much CMA (Circular Wire Area) you need based on the application.
This isn’t always an exact science as the cable length and speaker type will
change your calculated answer. The speaker cable becomes part of the cross-over
network in the speaker. The amplifier sees BOTH components as one load.
Since the cable is seen as part of the speaker, it
is easy to understand that the “reactive” relationship is between the speaker
plus speaker cable and the amplifier.
Speakers vary by design so the overall speaker component back EMF
portion of this load into amplifiers varies. Amplifiers of differing design
react to the back EMF and the overall performance can be hard to predict. The
goal is to “remove” the cable as best we can between the amplifier and speaker.
Cables should not be tone controls, but that’s the goal of EVERY component!
The analysis below looks at the calculations that
have been made to settle on the total CMA area for benign reactions to the
frequency response of a typical set of loudspeaker loads. And yes, these are
NOT real time resistive loads but as always, an approximation.
The general rule of thumb is that you want the total
speaker cable resistance to be less than 5% of the speaker impedance PLUS the
cable resistance value to avoid speaker frequency response interactions;
VOLTAGE DIVIDER FORMULA
Vout = Vin x R2 / (R2 + R1)
ICONOCLAST™ total CMA size mitigates appreciable calculated frequency response changes, and stopped at 9600 CMA (10 AWG).
For most practical applications of 0 to 35 feet,
9600 CMA per polarity should work well to be resistively invisible to the
speaker, or amplifier. We want the load to be the speaker, not the cable.
HOW we get to the approximate 9600 CMA per polarity
is the hard question. For those that want the easy way out we have one, 1313A.
If we want to see if we can DESIGN a better MEASURING cable let’s see what can
be done with Belden technology.
In order to figure out what best to do, I looked at
things that indicate what NOT to do. We all know by now, that multiple smaller
wires (to a point!) are better than one fat 9600 CMA solid or stranded
wire. The operative here is, to my ear,
TIME based issues at audio. You want the signal to be most uniform through the
wire for improved current coherence (more identical frequency arrival times).
To make that happen, we decrease the wire size so that the skin depth
penetration goes deeper into the wire, evening out the differences in current
magnitude with respect to frequency.
This technique better aligns the signal speeds through the wire. I said
“better” as there is no perfect way to do this. But we can certainly be better.
The depth is calculated based on frequency and material. The wire size does not
change the penetration, it DOES change the minimum current found in the center
of the wire. The smaller the wire, the closer the center current magnitude
matches the surface current as signal frequencies go up.
Studies were made on various geometries that would
hint at what type of conductor to use, and how many. What various design
limitations be “inside” the ~9600 CMA resistive box we want to be within?
Probably the easiest approximation for a cable with
multi-sized wires is a flat design. Yep,
line those wires up and stop when you reach the proper AWG size. The parallel wire tested issues lead (pun
there?) me away from this simple design. Why? I looked at our TEFLON® ribbon
cable for that answer.
A really nice “flat” TEFLON® Ribbon Cable
Above is a TEFLON® ribbon cable I used to test
polarity symmetry, and capacitive symmetry WITHIN each polarity. The two tables below graph the capacitance
from the outer edge wire to the opposite polarity, all opposite polarity wires
5.33pF (+/-2.59pF) @1 kHz
The data says that the CONSISTENCY of FLAT cable is not perfect. The closer each wire gets to the opposite polarity, the higher the capacitance. The GROUND reference is more robust the closer we get, and the less distance between two wires, all else the same, the higher the capacitance. We have EACH and every wire, for all intents and purposes, acting like a different wire. ANY cable with more than ONE wire per polarity will have this issue to contend with. How can we do better on capacitance control in each polarity?
For the answer to that we need to turn to inductance. When
you separate the two polarities in a flat design, inductance is seemingly well
controlled. Each parallel wire has current going in the exact same direction in
each polarity half so the magnetic fields CANCEL one another. The closer to the
inside polarity separation zone you go, the more the opposite polarity’s
different current direction upsets the SYMMETRY of the inductive cancellation
process. There is non-linearity through the “flat” polarity, too, but it is
worse near the edges of each polarity where the “design” changes.
Two wires with the SAME current direction next to each other
cancel some of the fields’ gauss density between them, and two wires next to
each other with opposite polarities reinforce the magnetic field lines.
Below are two close proximity wires. Notice that the current
direction “adds” between the wires with the magnetic field flux lines in the
same “reinforcing” direction. If we FLIP the current direction of one of the
wires, the currents cancel but now we have two of the same polarity to get the
cancellation effect. This is the problem with ZIP cord. We can get low
capacitance, but it is not practical to get the lowest inductance.
To prove a point, a single bonded pair used in ICONOCLAST
measured by itself is 12.5 pF/foot and 0.196 uH/foot inductance, about what
1313A reference zip cord is (chart below). This isn’t the best reactive
variable balance of L and C for a premium current delivery cable.
In the tested flat design there are inconsistent ground
plane issues that have to be resolved, AND there are inconsistent
electromagnetic field cancellation properties, too, through the “flat”. The
problems are locked-in by the geometry of this cable specimen, same as the
issues with zip-cord.
What is GOOD about a flat cable that and can we use those
positive attributes and mitigate the bad aspects? The answer to that question
lies in a BONDED pair used at RF frequencies. To get to the answer for speaker
cable, we need to re-invent what a BONDED pair does at audio. Re-designing a
bonded pair for audio leads to what size and count wires we can manage in
forward processes. We STILL don’t have the conductor size or quantity question
answered after all this.
What is a bonded pair? A BONDED pair is two co-joined wires.
A super geometrically consistent zip cord design with superior adjacent wire
BOND technology. The precision C-C of each wire controls impedance at RF to
incredibly small variation.
A zip cord removes a lot of symmetry complexity for poor
magnetic field cancellation properties. Adding wires to the zip cord to make it
a FLAT cable just adds to the capacitive and inductive “cable in a cable” issue
as every wire becomes its own drummer. Coherence is improved with more small
wires that add to the same CMA, but we don’t really have “one” like polarity
for each signal anymore.
Tests show the inconsistent capacitance in a FLAT
arrangement. Tests can also show the INDUCTANCE issues with zip- cords. A
single bonded pair is 0.196 uH/foot inductance. This value is far too high for
the state of the art R, L and C cable that is the intent of the project.
How is using another bonded pair zip cord component going to
fix this mess? The answer is in the XLR cable. We need to build STAR QUAD
arrangements of BONDED pairs! Visualize the currents using the right hand rule;
Like the XLR, two BONDED pairs in a QUAD arrangement show
ideal field cancellation with LIKE polarity current all in the same within the
same polarity. This field cancellation property of star quads tells us
fundamentally we need two polarities using many wires in a star quad
arrangement. There isn’t an answer as to how, yet, just that a true star quad
is a key element we need to keep.
The solution was a compromise, as is usually the case in audio
cables. The design devised a way to create star quads THROUGHOUT a process that
varied between near perfect, and slightly imperfect. It was done with 100%
consistency within each polarity so every wire measured the same inductance and
capacitance to the opposite polarity, and made significantly lowered inductance
with only a moderate rise in capacitance. The capacitance was increased on
purpose, I might add! More on why I did that later.
BONDED PAIR STAR QUAD ARRANGEMENTS IN PRACTICE
The above illustration shows the variation in the
STAR QUADS between like bonded pairs in a polarity. The question is does it
work; capacitance measured 45 pF/foot between polarity wires and inductance
measured 0.08 uH/foot. Capacitance variation, and the electromagnetically tied
inductance variation, is superb.
STAR QUAD POLARITY TESTS
The difference in reactive stability between each wire in a single polarity, and BETWEEN each polarity can measures significantly better in ICONOCLAST.
is +/- 0.5 pF @ 1 KHz or more than 5 times tighter variation than the 8R28064 flat cable.
was done was to BRAID, on a GHz capable braider, the needed wires to arrive at
the 9600 CMA DCR requirement. The braider needed a symmetrical arrangement so
an even number of bobbins was chosen, 12.
This is 24 wires per polarity. 9600 CMA / 24 = 400CMA per wire, or a
0.020” 24 AWG wire.
braid DESIGN is not forthcoming, so the balance of electricals has to be
understood. Several, several design
iterations were trialed before I froze the design around the proper braid
relationship to arrive at a suitably balance reactive cable measurement.
will “guess” that ICONOCLAST is a BONDED pair ETHERNET cable, and it is not.
The REASONS and the DESIGN are not the same at all. All that is the same is the
coincidence of a 24 AWG solid copper wire common to Ethernet.
polarity is BRAIDED and FLATTENED into a, you guesses it, FLAT shape! We
essentially “fold” the flat cable over on itself into ONE polarity. Then,
opposite polarities are tightly bound to keep LOOP area to a minimum, critical
to inductance as the formula is GEOMETRY controlled, not the dielectric.
BONDING OF TWO POLARITIES
Measured Rs (skin effect /
The nature of the magnetic fields can be indirectly MEASURED with an Rs
measurement. The flatter the Rs, the better the skin depth / proximity effect
are managed. Proximity effect is the currents in each polarity being “pulled”
to the inside edge of each conductor, and away from the outside edge. This
impacts conductor efficiency.
ASSEMBLY OF BONDED POLARITIES
An awful lot of testing was done to identify the weaknesses
of various designs. We wanted to avoid;
Inconsistent capacitance in each wire.
Inconsistent inductance in each wire.
Inconsistent ground plane interaction between wires and
wire DCR between all wires.
polarity DCR values (too high or low total CMA).
dielectric performance between each wire.
frequency coherence in each wire.
all the testing, a 20-mil wire diameter in a 24 wire (12 bonded pairs) woven
polarity was created to match the design to the electromagnetic
requirements. The final design that
drove the final wire size is 100% symmetrical in every measure on every wire.
single polarities achieve class leading performance in polarity-to-polarity and
wire-to-wire consistency while also providing exceptionally low reactive
variables. The superposition of the magnetic fields drive inductance down from
0.196 uH/foot to 0.08 uH/foot, a 59% reduction in inductance, while holding
capacitance to just 45 pF/foot. L and C can be CHANGED based on the woven
DESIGN, but was optimized for speaker cable applications.
TEFLON® was chosen as it is again, the best solid
dielectric there is. I needed a thin wall to bring the wires close together for
inductance reduction but capacitance is an issue with 24 closely spaced wires.
A capacitor is two parallel conductive plates with an insulator between them.
To lower capacitance, I wanted a low dielectric constant plastic, Teflon®. To
achieve the required low capacitance, more needs to be done to “thicken” the
insulation without increasing loop area effects.
This seems impossible to do, but it isn’t with the
woven design described above. The final insulation wall was driven by BALANCING
capacitive gains with inductive reduction.
Dielectric geometry allowed this balance to be accomplished.
The requirement to meet capacitance ALSO drove the
design to a weave pattern. Each polarity is SEPARATE from one another. There is
NO interweaving of same polarity wires.
Some will ask about wires with several AWG sizes.
Current will flow along the path of least resistance. This does not mean
current won’t flow in specific wires, just that the majority of the current magnitude
is shifted to the easier path. EVERY wire will have current at ALL frequencies.
The magnitude will change and follow ohm’s law. Many differing wires sizes and
electrical lengths can impact the signal arrival times across the audio band
based on physical conductor lengths in composite wire size designs.
If we take two wires with the same exact skin depth
(same frequency point being considered) but one wire has twice the surface
area, more current will flow into the larger surface area wire. It offers less
resistance. But, the lower resistance wire is a larger wire and isn’t what we
would like if the current across the wire is to be more uniform. Bigger wires
are better at lowering resistance at a given frequency because they have the
most surface area. We use this at RF with a “skin” of copper to carry the
lowest, yet still high, frequencies efficiently. The wire’s core under the
copper is a material that is “filler” and has no current flow: steel, aluminum,
At lower frequencies the current is diffusion
coupled evenly through the ENTIRE wire. So if you send JUST low frequencies,
use low a DCR wire as you can get.
Those are the extremes. Audio is weird in that we
need to improve current coherence through the wire while it is trying to MOVE
to the outside surface. We don’t care about attenuation as much at audio since
it is negligible. We make the conscious decision to go for forced current
coherence with more SMALL wires. This technically violates the practice of more
“surface” area for lower attenuation at high frequencies for current coherence.
Big wire is more surface area for attenuation while small wire is better
current coherence but higher attenuation.
If you use one wire (interconnect) the current delivery has to be
considered to the load. RCA and XLR cables have near zero current flow into the
high impedance load so we can go for signal current coherence and suffer little
attenuation. Speaker cables can’t use too few wires as there are 20-30 amps
coursing through a speaker cable.
Audio is trying to TIME align the low and high
frequencies, so the best, and most consistent, way to do this is to use more
small wires that add-up to the low frequency DCR needs, and are small enough to
FORCE the wire to see more and more cross sectional current usage at higher
frequencies. This means several small
insulated wire that all need to be the same “single” wire.
The unique woven design does a LOT to reduce
inductance and associated capacitance. How is 59% reduced inductance over a
single bonded pair achieved?
Star quad wire arrangement.
Allows ideal geometry for low field strength.
Boned pair like polarity wires.
Allows star quads to be formed throughout the weave.
Separate polarity halve fields are NOT parallel, reduce field
Fields between polarities have some cancellation (wires that cross at
ninety degrees cancel) since the cross at ANGLES, and not ever parallel.
Controlled Proximity effects / Skin effects
Measured Rs flat to 20 KHz.
Low dielectric constant plastic.
Thinnest possible C-C with the lowest cap.
Woven pattern averages out the wire-to-wire distances significantly.
Woven pattern separates the wires and “tricks” the bulk capacitive
value to be far lower.
The last point on the capacitive reduction is also
what we like in a FLAT design, but it is inconsistent. Average distance between
any two wires in a braided polarity and thus between polarities is far more
consistent. The weave moves all the
wires evenly, and consistently, to a closest proximity position and a max
proximity position throughout the weave.
Capacitance and inductance DO vary, but they are exactly the “same” wire
and at the same time as every other through the weave. The fattened weave holds
overall capacitance to an unexpectedly low value of 45 pF/foot in a cable with
such high conductor count.
Low inductance leverages the same current direction
in the bonded pair’s combined with the star quad wire geometry periodicity (end
view photo above). And finally, the TIGHT textile weave between polarity halves
force a low loop area and with wires never being parallel, further reducing
The overall reactance of the cable is shown in the
The chart illustrates a significant drop (yellow
trace) in cable impedance compared to 1313A (blue trace). We know all we need to know to figure out why
this happened. The velocity, although variable, is nearly the same at each
SPECIFIC swept frequency point. We need to look at frequency by frequency
calculations. The capacitance is linear across the entire audio band so that’s
a set value.
We have a set value of capacitance, and a nearly set
value of velocity (there will be slight variation) at a given frequency. What
is CHANGING is fundamentally the capacitance between cable designs for
The impedance equation is influenced by the change
in capacitance and thus lower measured impedance as the capacitance shows up in
the denominator of the impedance equation.
Increasing capacitance from ~16 pF/foot to ~45 pF/foot decreases
ICONOCLAST cable impedance. Speaker cables require low inductance and to get
there without shooting capacitance through the roof. DESIGN is the overriding
requirement, and materials alongside unprovable theory, are second.
Now we know why ICONOCLAST has the capacitance it
does, as I can balance the inductance to industry leading values AND keep cap
low, yet not so low as to increase impedance too high relative to the input
requirement (impossibly low speaker impedance 8-ohms ideal). Cables go UP in
impedance as you drop in frequency, the opposite of what we want. Listening
test have to decide if the superb inductance or impedance matching with much
higher cable capacitance is ideal. Quick
calculations will show capacitance problems with 8 ohm cables at audio once an
amplifier is attached.
Don’t ignore the reactive time constants of L and C.
We want an 8-ohm cable with NO L and C and zero resistance and you can’t do
that. Getting cable “impedance” reasonably low is more reliably safe for
amplifiers and TIME based distortions (lower L and C).
Shield material and design considerations.
I kept this topic here on purpose. Some may already know that low impedance
cables signal levels negate the need for a shield. And that’s a good thing
because a shield over a speaker cable is darn near ALWAYS a bad thing for two
shield will always increase capacitance of the cable. The question is how much.
mitigate the capacitance increase, the shield must be moved significantly AWAY
from the core polarities, increasing the size of the cable.
Shields are ONLY beneficial if the environment
demands them. Shields inhibit the performance of cable in most cases. Coaxial
cables being an exception as the shield defines the cable’s natural
IMPEDANCE. The ground plane proximity
and uniformity are vitally important with short wavelength RF cables. Coaxial
cables do just that. Audio is not RF, and these shields are more FUD devices
than actual benefits, especially in speaker cables that have signals orders of
magnitude over the background noise. Incidentally, the woven pattern in
ICONOCLAST has a built-in immunity to RF not that that RF immunity is evident
in the use of the cable.
View a SHIELD as a rain coat; great if you have
water flying around but a major hindrance if you don’t. Audio seldom needs
shielding on low impedance cables and here is why;
Magnetic fields decay rapidly with distance; ratio
of 1/x^3. The best defense is to MOVE the low frequency electromagnetic cables
away from one another. The foil and even braid shields are higher frequency
shields that are ineffective at much below 1 MHz. Magnetic fields lines need low permeability
shield material (something a magnet will stick to) to route flux lines away
from sensitive devices. A faraday cage is an example you can put something into
to do this. Low permeability metallic shields are a pain to use (stiff and
heavy). DISTANCE is the best remedy.
For EMI and RFI, the foil and braid shields used on
Interconnect cable will be fine for RFI ELECTRIC field issues, but NOT
20Hz-20KHz magnetic fields. Interconnect
cables MAY have wide band input op-amps that can be needlessly hampered by RFI
on the line. Speaker cable signal levels are many, many orders of magnitude
above the RF and ICONOCLAST speaker cables aren’t a good RF conductor due to
the weave pattern in the design.
Jacket design and material considerations.
All ICONOCLAST cables use FEP as the jacket to
reduce UV sensitivity, plasticizer migration and chemical resistance. The cables are designed to last decades.
SUMMARY – Little has been left to chance in the design of ICONOCLAST cable. All the products are born from strict measurements and the management of known electrical parameters. Belden’s philosophy is to make as low and R, L and C cables as technically capable. The improvement to some may be unimportant. To others, and using different systems, they can be significant. The closer we manage the knowns, the better the tertiary elements will move along with those improvements. All cables “react” differently. ICONOCLAST is designed to offer the most benign interaction possible between your amplifier and speaker by leveraging high speed digital design principles to the much more complex audio band.
I have four 1 meter XLR sets going from the bottom component, up to the pre amplifier component. The natural LOOP needs to be allowed to form, unobstructed from the rear wall. This takes about 12″-13″ of clearance from the back of the unit to the wall. The rack may be closer to the wall than 13″ based on the depth of the rack. If the rack shelf is 24″, and the device is 19″, the rack will be only 13″minus 5″ or 8″from the rear wall, for instance.
Space the racks and devices APART helps, too, so a 180 degree loop back for one device on top of another is too severe. A bend radius of 8″ or so is needed (see the picture). If devices are such that the XLR are well offset, a piggy-back stack can work. The distance BETWEEN the XLR jacks is what is important, up and down or left and right or a combination of the two works.
The XLR will twist no more than 180 degrees to plug in for ANY wire path. With a loop-back, a change in the vertical alignment, to the left or right, will force a TWIST in the cable. Same with a straight through if the equipment is shifted left or right. Again, this is simply how XLR works.
Plug the MALE XLR in first. The female has a thumb tab that lets you quickly see the direction to go that puts the least torque into the cable. Simply rotate the female XLR to snap it in. All polarized plugs work this way. The BAV is designed to simply have a very low bending moment design that absorbs the exact same twist, but with less force…thus it can take severe routing.
ICONOCLAST used with adequate rear clearance can use 1 meter cables, but the FULL natural loop needs to form to allow the torque to be absorbed along the full length of the cable. This is worst case as the jacks are pretty much one straight above the others.
ICONOCLAST are true AIR tube with the best dielectrics available. These cables will require proper routing and rack positions. For those that can not conform to the reality of the design, I have engineered the BAV to be the best flexing (first requirement) and best performing for the flexibility. These will accept harsh routing.
NOTE: This paper was originally written prior to the introduction of Iconoclast Gen2 interconnects, so while some references are to the future, that future is now….
BACKGROUND: To possibly improve the performance of
the XLR, to maybe achieve even lower L and C,
we would need to revise the current design…and it will jump up the
electromagnetic complexity. The balanced of L and C would shift some but the
coherence will improve substantially.
Changing the “conductor” to a four insulated wire structure
will lower INDUCTANCE through signal phase cancellation. The star quad
arrangement will retain CMRR for NOISE reduction. Four smaller wires will
improve PHASE, and lower wire loop DCR to mitigate ground loops.
PROTOTYPE IMPROVED DESIGN:
Capacitance is the DISTANCE between
the plates (wires) and dielectric material(s).
Inductance is two-fold;
The electromagnetic field cancellation.
The loop area between the wires changes inductance.
For inductance dielectric doesn’t really matter,
inductance is DISTANCE.
We will have the same nearly loop
area in the design (C-C distance is the same) but each conductor in the new
design will further remove signal electromagnetic fields based on the
cancellation geometry. Inductance should drop compared to the single wire
The capacitance requires the same
meticulous attention paid to as the group dielectric. Since we are keeping the
cable the same size so the capacitance HAS TO go up as we have more wires
parallel to a dielectric (the center X-filler, beading and outer tube) , and
closer to the dielectrics. The conductor size and dielectric determine the final
size. The added wires and X-filler close to the wires are the main contributors
to the required capacitance increase. But, lower inductance improves PHASE
shift, and your ear is most sensitive to.
The current coherence, the main
objective of the design with minimal L and C changes, is based on the skin
depth penetration changes going from 1 x 0.018” wire to 4 x 0.010” wire for
4 wire “conductor”
Technically, four -wire per
conductor will increase capacitance some as we have more wires parallel to a
dielectric, but the current coherence
improves substantially, time aligning the low to high frequencies.
18214.4 μ inches = 18.2 mils @ one skin depth.
One skin depth is defined as when the surface current is 37% smaller going into the wire. If we had a wire that was 18.2 mils in size, the CENTER of the wire would have only 37% of the current measured on its surface.
Skin depth equation (below) is a
squared equation, so removing wire depth rapidly increases the inner current
magnitude. Dropping from 20 mils to 10 mils is a 4X improvement in current
The very good,
and easier to make, current design does NOT use electromagnetic signal field
reduction technology I developed for the speaker cables in the series 1 signal
leads. The current XLR design relies on reduced loop area and uses AIR to
reduce the capacitance to a minimum for a given tighter spacing to achieve
inductance. The better the dielectric
the CLOSER I can physically locate the signal wires for a given capacitance,
thus lowering Inductance. The size of the wires determines the current
coherence, and with more uniform effect of the dielectric around each wire with
respect to frequency. The smaller the wire, the more uniform the velocity of
propagation from low to high frequencies.
An XLR cable’s
external noise utilizes CMRR based on all four noise signals being equal on
each wire and which cancels those noise signals in a star quad design through
electromagnetic field cancellation. If we look at the four wires, and using the
right hand rule (current out of the page). All the external noise currents in
the wire go CCW around each wire suspended in space. All the electromagnetic
fields cancel adjacent to any wire and across from any other wire. All the
fields superimposed onto one another forming a nearly ideal cancellation
circuit. Nearly perfect because stray magnetic fields would extends OUTSIDE the
four wires and reinforces the field. A first approximation says that this
doesn’t happen. The stronger fields are closets to the wire and cancel most
aggressively. Theoretical outer fields are weak, and don’t reinforce nearly as
much as the inner fields cancel.
TWO WIRE FIELD CANCELLATION ASSUMING FIELDS EXTEND PAST THE CENTER BOUNDARY. MAGNETIC FIELDS WILL CONCENTRATE BETWEEN THE TWO WIRES, HOWEVER,AND CANCEL:
OVERALL 1 X 4 WIRE CONDUCTORS ELECTROMAGNETIC SIGNAL FIELD CANCELLATION:
We DO NOT see
this nearly “perfect” rejection of signal magnetic fields to reduce the
inductance in the signal fields for series 1 RCA or XLR cable. We have a PLUS
and MINUS balanced signal current direction whose fields are only partially
cancelled. The partial field partial cancellation RAISES the inductance above
“zero” theoretically as we have a stronger field, and separated by the distance
needed to lower capacitance with any dielectric. The old design has ~36% higher
inductance, and thus worse PHASE shift than series II (0.015 uh/foot is reduced
to 0.11 uh/foot nominal). Lowering inductance directly lowers phase. See the
QED phase analysis measurements on a variety of cable;
QED – The Sound
of Science www.qed.co.uk/downloads/qed/soundofscience.pdf
OVERALL 4 x 1 wire XLR CMRR INTERNAL (signal energy) ELECTROMAGNETIC FIELD CANCELLATION:
The two MINUS
fields cancel between themselves.
The two PLUS
fields cancel between themselves.
But a MINUS to
PLUS field REINFORCES the overall magnetic field.
makes the field stronger and the loop area effect worse.
BODY –To make improvements, we need to
reduce the signal electromagnetic field to ZERO, in theory, both from an
external interference view AND an internal electromagnetic conductor view. To
do this, we need to BALANCE the music signal by SPLITTING each of the four
SIGNAL wires into FOUR, or sixteen separate wires.
critical change will theoretically remove the signal field currents that
interact with the loop, creating inductance. It will also significantly improve
the dielectric group and Phase delay by forcing the dielectric to be seen more
uniformly across the 20-20KHz frequency range with smaller wires.
capacitance low for a given loop area, we need to use AIR around the wires, and
to make sure any plastics that touch the wire are super low dielectric constant
materials (FEP mini X-filler and external FEP bead wire). This is why the wires have to be BARE copper
with NO insulation around them. Only the
tangential surface of the FEP filler and FEP beading, the rest is air.
Capacitance is dielectric AND distance related where Inductance is distance and
electromagnetic field strength.
Each of the four
wire will be shorted together to make the typical four wires in a star quad.
The wires are 10-mil diameter 30 AWG for a total CMA of 4 x 102 =
400 CMA. I used 4 x 0.018” in iconoclast for a total 18 x 18 = 324 CMA for each
signal wire. 400 CMA is slightly lower DCR than the current design improving
attenuation and mitigated ground loop voltages.
filler is foam FEP to manage capacitance. The power carrying signal braid
should also be as far away as possible from the internal signal wire QUAD
structure to lower the ground plane inside the cable, lowering capacitance.
This means making the outer belting thickness under the braid to the best fit
for an XLR connector, but not too big as the reduction in capacitance is a squared
law variable, once a threshold is reached, more is not too beneficial.
The X-filler is FEP, as would the
30-mil beading wrapped around the QUAD wire to lower the dielectric nearest the
wire where it is most critical. The material issues all control capacitance,
The overall belt is solid FEP, with
a 36 AWG BC braid and a drain wire. A final solid FEP jacket finished the
0.230” BELTING 0.030” BRAID 0.015” JACKET 0.030” TOTAL 0.305”
Does it really work on initial Capacitance and Inductance measurements? The final design using the ICONOCLAST™ all FEP design for ultimate performance appraisals measured as follows:
Lab Rqst-177575 Sample ID – 60156Y (PDC2842)
Requestor – Galen Gareis Report
Generation Date – 22 June 2017
Capacitance @ 1 kHz
per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
Meas: 18.23 pF/ft
Inductance @ 1 kHz
per ELP 424, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
Meas: 0.10 µH/ft
Lab Rqst – 177587 Sample
ID – PDC2842
Requestor – Galen
Gareis Report Generation Date – 29 June 2017
Capacitance @ 1 kHz
per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
Cap @ 1 kHz
Spec: 10.5 pF/ft max
Meas: 17.48 pF/ft
Inductance @ 1 kHz
per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
Meas: 0.10 µH/ft
Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument
Control Software with Velocity of Propagation program and a GPIB card
1×4 Ref. Design
231 pF/12.67′ = 18.23pF/ft
1.29 μH/12.67′ = 0.10 μH/ft
To really get better XLR
performance, both loop area and the field cancellation technology need to be
leveraged, with the latter being most critical. The capacitance is all about
materials and DISTANCE between them. Improving inductive field cancellation has
the added, and significant, benefit of improving signal coherence through four
smaller wires and phase with lower inductance while improving attenuation
A cable with smaller signal wires
and better coherence, low inductance (better phase) and slightly higher
capacitance will sound better sounding than a cable with larger signal wire and
less coherence, higher inductance and lower capacitance…as long as capacitance
isn’t too high!
The prototype run does indeed lower inductance with the
expected rise in capacitance.
0.15 uH/foot (X) + X = 0.1 uH/foot, X= 33% lower. Or the current design is 50% higher.
12.5 pF/foot (X) + X = 18.23 pF/foot = 45.8% higher Or the current design is 31.4% lower.
Since the original design is working from such low L and C
numbers, the percentages are not really illustrating the advantages of the
improved signal coherence with much smaller wires, and an advantage that should
play out in audible performance. The
-3dB first order filter frequency is still well above the audio band so first
order filter phase distortion is not going to be an issue. What must be the
major contributor is coherence with the smaller wires. Rs response, while
lower, is hard to quantify.
Rs (swept frequency
The 4×4 XLR lowers swept Rs (proximity effect) values
significantly, and flattens the high-end linearity. Can you HEAR that
improvement, over the single the wire design? The truth is BOTH are
superimposed when the wire is used, and pushing the XLR designs to as near
perfection is certainly a better and better design. The lower DCR is evident in
the trace compared to the 1×4 25 AWG wire as is the flatter upper frequency
The RCA interconnect has also been updated with the new 1×4
(ONE wire made with FOUR conductors) design. The reactive variables will track
with frequency like the single wire designs, but map to the altered L and C
The following table shows the effects of changing the wire
size and number. The 4 x 4 has almost the same CMA as a single 22 AWG, but 1.82
times more total circumference, which shows up only at increased frequencies.
The lowest frequencies are essentially DCR.
The maximum Rs is lower with
the 4×4 design. Beta test feedback from customers on the 4×4 has been extremely
positive and, consistent with the numbers, shows this revision to be a
significant upgrade from the original Iconoclast design for analog
DCR INTERCONNECT LOOP CONSISTENCY
The interconnect cables of a given wire design (single to single and
verses quad to quad) have essentially the same loop DCR values.
From the Rs chart above at DC,
4×1 XLR and 1×1 RCA are 34.11
and 39.19 Milli-ohms/foot respectively.
4×4 XLR and 1×4 RCA are 27.53
and 28.79 Milli-ohms/foot respectively.
How was this done? The double
braid on the RCA was necessary to mitigate ground loop DCR variation between
sources, and the DCR was designed to be near a “free” return path for loop[
DCR. The loop resistance ios the braid plus the
conductor. But, the braid DCR is so low that the loop DCR is pretty much
the RCA center conductor. This is true for eother design.
The XLR DOUBLES the number of
conductors in each leg as a star quad. This reduces the DCR to one-half the
conductor’s value. Thus the two pairs in parallel are the same DCR as a single
This was also done on purpose
to make sure that the RCA’s loop performance was as good as the XLR, and that
the RCA BRAID was essentially a ZERO DCR return path between grounds. If the
RCA braid was insufficient DCR, we would see more divergence between the two
singel ended and balanced design.
The measured XLR electricals are very good, and follow
design theory perfectly. ICONOCLAST once again shows that proper engineering
fundamentals are paramount to performance. Sound Design Creates Sound
1.0 Conductors 1.1 Copper Size 2.0 Dielectric Materials 3.0 Dielectric geometry 4.0 Shield Material and design considerations 5.0 Jacket design and material considerations
BOTH of the copper conductor
and size considerations were answered when we started the RCA cable. We don’t
want to change the current coherence with a differing conductor diameter if we
are to mirror the reactive variables, too. We need the same exact wire to
shield reactive L and C parameters in each cable in the end configuration
design. The geometry of each cable is entirely different so how to you do that?
That is, assuming you want to match
the RCA and XLR properties and maintain the same signal quality…and we
certainly do. There is no reason to copy
a bad sounding RCA cable when designing an XLR, so the RCA is designed FIRST.
One difference in the
XLR is that we are going to use FOUR wires in a star quad configuration. (Note:
in our “Gen 2” XLR product, there are sixteen wires — four wires
making a star quad in place of each single wire in the design shown below. For more detail, see the last paper in this series.) Four wire XLR cables use two cross-connected
wires for each polarity, which doubles-up the wire AWG for lower attenuation.
Two 25 AWG have the DCR of a single 22 AWG yet has way better signal coherence
by using smaller wire.
could have used a cheaper and easier two wire XLR design but the inductive and
signal coherence benefits of a star quad are too good to pass-up. If I can get
the materials and quad design to achieve a high enough level of performance it
is a better cable design.
quads have a higher degree of CMRR (Common Mode Rejection Ratio) when properly
signal balanced. There are three primary reasons for this;
The two or four wire stranding “twist”.
The differential encoding.
The outer shield properties, but only at RF frequencies.
wires of a star quad are a “positive” voltage, and two wires are a “negative”
voltage (180 degrees out of phase), hence the term “balanced”. If the cable were a teeter-totter, it would
sit level. Some call this differential mode since each signal is equal but
Differential Mode Transmission
Perfect Wire Balance Equals Less Noise
In the example above we show two
wires, but the system is the same in a star quad. The signal we WANT is encoded
as +2 volts and -2 volts. The noise can’t “change its spots” relative the
cable’s twisted pairs and shows up as the same voltage on each wire, +1V noise
in this example. The TWIST ratio helps make sure that the wires see the noise
the same amount of time and this is vital to the function of the circuit.
Here is where the balance is so
important; the signal IDEALLY becomes the superposition of ALL the voltages, or
+3 volts and -1 volt. No more, no less. The signal voltages are STILL exactly 4
volts “apart” from each other; +2 to -2 with no noise and +3 to -1 volts with
the noise. The signals are fed into a
difference amplifier that, you guessed it, looks at the “difference” between
the two voltages and see’s 4 volts with, or without, the noise. The noise is
absent in a perfect world at the difference amplifier’s output.
In order to do this, every wire
has to be presented to the noise in the exact same way via the cable twist and
has to be the same length so the signal stays TIME aligned down the wire and
has to have the same attenuation. The difference amplifiers need to be nulled
perfectly between gain halves. Believe it or not, this gets done really well
with good quality products.
The control tolerance of the
copper is 0.0005”, so attenuation issues are mitigated and CUB (Capacitance
UnBalance) tests insure we see MIL standard quality in the finished cable. All
quality types of copper can be used in the XLR design. It is the overall
structure that is the most “magic” and not as much the copper itself, although
the copper draw process does influence the sound.
We have several variables that
aren’t present in a coaxial cable design to contend with;
CUB, Capacitance Unbalance or, each wire shows a
differing capacitance to ground.
DCR unbalance, each wire has to be the same DCR.
CMRR remainder, the differential signs have to NULL to
the exact same point neither above nor below reference ground.
Lots of words, time for a picture;
ICONOCLAST™ XLR END VIEW
The above CAD drawing is what we have
inside our XLR design so far (well, I ignored two wires in the drawing).
Remember I wanted to make L and C
reactive variables EXACTLY the same for each cable with EXACTLY the same wire
size and draw science? What else do we know? I also said that CAPACITANCE is
sensitive to the distance to a conductive plate area, and that means ALL the
way around the wire. The coaxial cable is easy; we purposefully put a ground
around the wire at a known distance that defines the capacitance ground plane
reference distance and inductive loop area.
In the coaxial cable, the center of
the wire to the inside of the tube is 0.098” / 2 = .049”. Ok, so what? This is what. The capacitance is
a squared law property and predominantly sees the ground closest to the wire.
The shield on the opposite side of the XLR cable, to a first approximation,
falls a way. We actually measure the capacitance BETWEEN the two cross paired
wires but the ground location still influences the capacitance. Also, we have
four wires that are capacitors.
This doesn’t “sound” good, does it?
We have four times as many wires and all have capacitance. Somehow this is
supposed to come out around 12 pF/foot (with connectors), same as the RCA!
Now for the inductance part, L.
Inductance is loop area defined. It could care less about the dielectric, but
the graph above shows a HUGE ~0.170” loop area! How is THAT going to get to the
0.15 uH/foot inductance of the coaxial cable? I could make something up, but
that isn’t as neat as what’s really going on.
To get capacitance as low as I need
it to be to match the coaxial cables, I use DISTANCE between the wires. And
yes, this DIRECTLY sets what the inductance will do…hold on a minute. By using AIR, I can set the C-C of the wires
to meet my capacitance target needed for the final tested value with two cross
wires connected and tested between them. AIR lessens this distance for a given
value of C so I can also manage inductance now. For inductance, L, the smaller
the wire loop area the better for a given value of total capacitance. Air gets
me far closer than any other dielectric.
How much air? Well, EXACTLY the same
as the coaxial cables! How do we do that? The standard answer is, “very
carefully”. Let’s look at a drawing;
ICONOCLAST™ XLR CHAMBER VOLUME
Still, so what? Yep, I agree, until
we compare this area to the area in in the RCA cable air dielectric; 0.00754 in2.
OK it isn’t exact; I missed by ~0.000009” in2. I use the exact same thread design around
each identical wire so it’s all the same area in the chamber as in the RCA.
Let’s do some reality checking as to
what it SHOULD be based on MEASUREMENTS and calculations.
We have the EXACT (can I say that as close as it is?)
same velocity of propagation based on the composite (air and plastic inside the
ground plane) dielectric; 87% at RF reference.
I measured the IMPEDANCE at RF @ 100 ohms, same as the
The dielectric constant can be calculated and from that
the VP, VP = 1/ SQRT (E).
And from that
composite dielectric I also know what the capacitance has to be.
Capacitance (remember that chart on
dielectric value and capacitance earlier?) is directly linked to the group
dielectric constant. I know VP, and I know the impedance, so I can calculate
the capacitance and then get the dielectric constant from that.
101670 / (C * 87) = 100 ohm
C = 11.68 pF/foot.
What does the cable actually measure
on capacitance? The chart below shows 11.767 pF/foot. Notice that the
capacitance values between each of any two wires has to be ~ 5 pF/foot to
“double-up” the two wires capacitance and still to arrive at a final ~11
pF/foot! Yep, that’s LOW capacitance. Capacitance adds in parallel so this is a
significant issue when a design uses four wires.
Below is the measured and calculated
imbalance of the capacitance between 1-3 and 2-4 cross wires’ conductors as a
“pair”; 2.02% unbalance, very low.
We seem to have the capacitance and
VP looking much like the coaxial cable. Remember, measurements include ALL the
approximations in the soup.
So what about inductance with that
WAY larger loop area? Isn’t that going to really kill this thing? No, because of some properties of magnetic
fields. Magnetic fields CANCEL if they see each other in OPPOSITE directions.
Inductance is the “reactance” or “resistance” to instantaneous flow of current.
If we can REDUCE the magnetic field lines, we can directly reduce the measured
We also know from the basic equations
that DISTANCE between the two wires is important. Keeping BOTH distance and
magnetic field line magnitude small lowers inductance, and removes the noise.
The picture below shows what’s going on…sort
of. For now, we’ll pretend the field’s ONLY go “inwards”, or inside the wire,
and stop there (they don’t). If the lines that extend outside each wire do the
OPPOSITE as the field INSIDE the wires, they reinforce the field! It is generally accepted that the flux lines
concentrate substantially BETWEEN the wires.
If we draw arrows that represent the
DIRECTION of the circumferential magnetic field waves AROUND each wire we get
what is shown below for a NOISE signal hitting the wire. We have TWO different
voltage polarities so we have TWO different current directions for the SIGNAL,
but the NOISE is the SAME direction in all the wires.
If you grasp a wire with all four of
your fingers, and point your right hand THUMB in the CURRENT direction, your
fingers will point in the field’s circumferential direction around each wire.
The arrows are a “part” of the noise current field lines “inside” the four-wire
NOISE FIELDS (all the same direction)
Where the arrows are OPPOSITE each
other in direction between any two wires, the field lines cancel. For NOISE
every field theoretically cancels. ADJACENT or ACROSS from any TWO wires we
induce field cancellation with a star quad design.
For the SIGNAL, we now have TWO equal
but opposite current directions.
This allows larger wire-to-wire
spacing in order to lower capacitance and also keeps inductance low. Inductance
is managed with field line cancellation geometry.
Now we know why I didn’t use a
two-wire system, you can’t manage CMRR.
Let’s look at the situation for the signal. Below is a simple picture of the field cancellation between four wires with opposite polarities wired as a star quad.
SIGNAL FIELDS (opposite directions)
Minus = Current INTO the page (CW rotation)
Plus = Current OUT of the page (CCW rotation)
What do we see? The all the signal
field lines DO NOT cancel. Adjacent wires reinforce, and opposites wires
cancel. Reducing loop area is the best way to manage inductance because we
can’t cancel all of the field lines, only some of them. This theoretical field
relationship limits the ability to reduce capacitance for a given inductance.
Using low dielectric constant materials to lower capacitance (Air!) allows
closer spacing needed for low inductance.
Is there a design that can, in theory
do BOTH, reduce signal and noise fields to “zero”? Within the limits of DESIGN,
yes there is. The ICONOCLAST series II
reduces both noise and signal field cancellation. The wires, in practice aren’t
EXCATLY the same distance apart and EXACTLY the same resistance, so we say “in
theory”. But, reducing the nose to 1000 or more times less and reducing the
inductance 27% is indeed achievable.
So, after all that explaining, how
does the star quad ICONOCLAST cable measure up? Tests at 1 KHz show the
following values below. The inductance between the two cross wire pairs of the
star quad are 0.15 uH/foot inductance…same as the RCA.
So what does the “reactive”
picture look like comparing the RCA and XLR? How close are they to being the
same? This swept test is the real deal. There are no approximations to fudge.
What we see above is impedance /
phase for the XLR and RCA superimposed one on top the other. Note that there
are four separate lines. We have two identical cables with exceptionally low
Shield material and design considerations.
yet one last thing to consider in the XLR design; the outer shield. A 95% BC
(Bare Copper) braid is used. Audio cables are not RF designs, and the braid
shield will NOT shield low frequency magnetic interference. The CMRR of the XLR
is going to do that for us. Excellent CUB, DCR unbalance and twist ratio all
aid CMRR. The braid DOES knock down RFI by 80 dB, so that’s a given. The shield
isolation @ RF mitigates NULL balance at high frequencies only.
or not, 20-20K is a predominantly magnetic field frequency range where the
B-fields decay at a ratio of 1/x^3. DISTANCE is the best solution for isolation
of cables with magnetic properties.
5.0 Jacket design and material
ICONOCLAST cables use FEP as the jacket to reduce UV sensitivity, plasticizer
migration and provide chemical resistance.
The cables are designed to last decades.
I hope that this design summary of ICONOCLAST RCA and XLR
interconnect cables shows how important good design is for ALL your audio
cables, and that every manufacturer has to manage all the same variables to
produce these results. There is little “magic” in the design of good cables.
There are indeed tertiary variables that we can’t measure, but those should not
influence the ones we can measure, or at least not excessively so. Mother
Nature abhors complexity, so the better managed the known variables in a cable
are, the more properly it may highlight “unknowns.” To put it another way, the more we put knowns
into their proper place, the better we may distinguish the effects of the
unknown. Wire draw science, for instance, can be heard better, and more fairly,
in a superior electromagnetic design.
Belden appreciates your interest in how quality
interconnects are made, and how / why ICONOCLAST RCA and XLR cables were
physically derived as you see them in their production form. We have no special
sauce or magic in our products, and I think that the cables perform as well as
they do BECAUSE we did not design around “unknowns” and then make it appear as
though we had unique influence on those unknowns in the design.
Truly low R, L and C cables are difficult to make when
consideration is given to all three variables to manage them in a truly
balanced fashion. The designs can be frustratingly simple looking but hard to
manufacture, as processes are pushed to the limits of current capabilities.
Belden’s focus is to make real measured values as low, and properly balanced,
as we can. ICONOCLAST interconnects represent the pinnacle of low frequency
measurements and electrical balance between the RCA and XLR (same
The next design analysis will look at the SPEAKER cable.
In a previous paper I
covered several issues that create signal distortion in audio cables. The most
demanding variables involve the TIME related distortions that the ear is most
sensitive to. Consideration must be made during cable design to mitigate the
TIME based issues through the audio band. The following paper is the journey
through the design process to arrive at a satisfactory RCA and XLR cable
design. I must stress ALL quality cable designers have to work with the exact
same known variables to solve problems at audio. Every cable is a compromise of
some sort as distortions can’t be eliminated.
ICONOCLAST has made the outlined design decisions to arrive at, what we
think, is an industry leading design based on real measurements.
DESIGNS CREATE SOUND PERFORMANCE™
RCA DESIGN BRIEF
1.0 Conductors 1.1 Copper Size 2.0 Dielectric Material(s) 3.0 Dielectric geometry 4.0 Shield material and design considerations 5.0 Jacket design and material considerations
The design process will start with the RCA cable as
this provides the most pristine electromagnetic properties possible due to the
seemingly simplistic design. Once all is
said and done it is “simple” looking. The more complex XLR will have to, somehow,
match the RCA’s electromagnetic properties if it is to be an “equal” on
measured attributes. If the RCA isn’t any good, I may as well start over again!
“…granular building blocks in copper can never fit together perfectly, but are rotated causing an unexpected level of misalignment and surface roughness. This behavior, which was previously undetected, applies to many materials beyond copper and will have important implications for how materials are used and designed in the future…”
battle for material supremacy continues. However, what we tend to discount is
that while the overall design of the tire we put on the car is important, the
rest of the car has more to do with what that tire does than just the tire. We
over spec the tire and vastly under spec the car. I’m intent on building the
car, not the tire.
decision to use copper is based on several factors, none of which were price.
Copper offers the best material for affordable cables with a significant level
of performance in more ideal electromagnetic designs. Far more expensive
materials in lesser designs won’t work, and far more expensive materials in
superior designs won’t work…for most of us anyway.
is available in several process treatments and after process treatments;
(as good as what used to be OF grade)
OFE (differing process, but far from vastly lower impurities content)
OCC (what is often called long grain type, and again a differing process).
treatments (used to improve copper’s PHYSICAL properties)
direction (music is AC. Which polarity do you like first and at what
I don’t use wire “quality factor” as
a design element since every contemporary draw science wire is of vastly better
quality than ever. Sure, some processes are more $$$ but there is scant
repeatable measurement that I can do other than conductivity, a passive
resistive measure that will influence R, L and C. The conductor type is an option
for the customer to listen to, only. There are differences. Belden just isn’t
in the position to create a pet project to define what isn’t yet scientifically
defined. That’s not our thing.
offers the three fundamental copper grades; ETPC, OF and UP OCC, as they DO
sound different in the exact same electromagnetic R, L and C referenced design.
No changes other than the copper, so we know what the culprit is. What we don’t
know, is WHY it is the culprit. Instead of making up a big old story, again, about
the material, we don’t. It is what it is in use and we leave it that way.
we don’t offer is what I can’t hear as a designer. Sorry, but I’ve yet to hear
CRYO treatments, intended to improve the wire’s PHYSICAL strength or grain
direction, change the sound. As far as grain direction goes, you can flip the
leads in any direction you want, as the wire’s grains all go the same way due
to the manufacturing process that we use. If you can hear the direction switch,
flip them any way you like. We won’t send you a bill for that!
material used in a superior design SHOULD sound as good as it can, and cost
isn’t a direct line to better sound. I ignored cost when I designed
ICONOCLAST™, either high or low. If my system didn’t allow me to hear it, I
didn’t use it (materials) or do it (process / design).
isn’t a paper on conductors, although I may have some things to say about
alternatives to copper them later on in another paper based on some
measurements and calculations I’ve done. We’re talking copper in this paper as
it is the very best economical solution that we have right now.
has a very low DCR, a reasonably deep skin depth to manage current coherence,
is pretty high in tensile strength for processing, and in most applications
resists severe oxidation. The grain structure is clearly visible in form, but
that alone is NOT what makes the different grades sound different. It is a
trait of the draw science, but does not have as much effect on the sound as you would be lead to believe.
solid or stranded wire? This, at least,
is easy. Is stranded better for the way the cable is used? Is stranded more, or
less, expensive? Is stranded easier or harder to process? Is the termination of
the cable better or worse with stranded wire versus solid? Are any gremlins
that I call tertiary variables (stuff there isn’t a measurement or calculation
for) removed if the truly measureable variables are accounted for between
stranded and solid?
– Solid wire wins hands down for this application. Every question is in solid
wire’s court. End use, costs less, processing cost, ease of termination and
lack of tertiary elements (all those diode effect “arguments” between strands
that, though, a note: the first generation of Iconoclast interconnects use
single solid wires for the signal-carrying conductors and that’s what’s
discussed in this paper. Our second
generation product (suitable for analog but not for digital due to impedance
issues) uses a star-quad arrangement of four separate wires, placed around a
separator, in place of each of these conductors for improved inductance; for
details see the fourth paper in this series.
Other than this change in the signal conductors, the “Gen 1” and
“Gen 2” interconnects are the same.
1.1 Copper Size / RCA
now have SOLID copper wire. The size selected sets the foundation for the whole
thing if we consider that the cable’s structure is supposed to allow a
conductor to be as near zero R, L and C measurement cable as we can design.
can’t use a conductor you can’t process. For the RCA cable, we want as small a
wire as we can process as this will force the best current coherence through
the wire (same current magnitude at all frequencies). The exact skin depth
calculation is a tool we use to gain the knowledge
to reduce the wire size in audio cables. At RF, we use it to tell us how much
copper to put over a STEEL support structure to maximize RF attenuation. Audio
is not RF, and the ENTIRE wire is used to move the signal and at ALL frequencies
concurrently, not the same issue at all in RF cable design.
RCA cables terminate into a theoretically infinite (47K-120K or there about) input resistor. We say impedance, but it is really as resistive as it can be made at the input op-amp level. Yes, purists will point out that input impedance DROPS some at higher frequencies.
the impedance is so high and the current is so low (it looks like an open
circuit) just use as small a wire as you can! Well, yes and no. It has to be
reliably terminated and secure in the end product, and it has to process evenly
under tension and not fracture from surface issues.
review of the end of process design backs into the initial design requirement.
Calculations and testing selected a 0.0176” diameter wire for ICONOCLAST. The
process has to handle less than 4-3/8 pound tension to avoid permanent wire
stretching. Wire was tested for the process requirement.
0.0176” diameter wire (0.0088” radius) is one half the diameters necessary for
one full 18-mil skin depth at audio, so we have significantly improved current
coherence through the wire @ 0.0176” diameter wire. Skin depth is FREQUENCY
driven for a given material. The smaller the wire the larger the inner current
magnitude will be relative to the surface current. We want as good a shot of
that as we can get.
RCA cable’s loop DCR will essentially be the center conductor in an RCA, if it
is made right, and ICONOCLAST is. The center wire governs attenuation. The
outer conductor is, in theory, infinitely low impedance so it nearly drops out
of the loop DCR calculation and leaves the center wire. The length of the cable relative to the input
impedance allows a SMALL wire at audio. At least attenuation works in our favor
at audio as it is a LOG relationship and gets really high very quickly as you
go up in frequency. For audio, we can relax a bit on attenuation as it is low
for the lengths we use and is in the right frequency range to stay low.
Attenuation is a passive “distortion” and is VERY hard to hear over TIME based
already made a critical choice in our cable. The wire material and size. We’ve
used good engineering practice to KNOW what the decision will yield. Now, how
to RETAIN all that the material / size wire can provide? That’s easy, just
stick it in air and find an infinitely low ground potential for our unbalanced
/ single ended wire!
this IDEA is easy. The execution isn’t. I don’t care about speed of the process
and / or costs as I’ve used REASONABLY affordable material as my conductor. We
can always go back and break the bank on conductor materials. AIR is free, but
expensive to get. Air is by far the best dielectric to have, and especially
nearest the wire were the influences are the worst on group delay. The closer
to the wire the dielectric is, the more it influences the overall velocity of
the composite structure (wire / beading/ then plastic tube thickness / then
decided to go the tough route and use air. We can use RF as a HINT at what to
do overall. We have used designs called semi-solid core dielectric RF cables.
These partially suspend a wire in a tube with a spirally wrapped thread. The
problem is that the wire SIZE and the core tube properties aren’t suitable for
audio frequencies. Even the choice of materials isn’t as important at RF as we
can reach a set impedance vector (real + the reactive inductive and capacitive
parts all added together) by tweaking the thread and tube dimensions.
3.0 Dielectric geometry
audio signal is very sensitive to the dielectric effects of the plastics near
it. I chose a specially made beading thread to get the job done.
above picture beading around the wire is a glass thread coated in pure
TEFLON®. I use a ROUND beading shape
versus square, as it touches the wire at the tangent points for the very LEAST
effect nearest the wire. The electromagnetic field sees the entire cross
section of the plastics and material between the wire and the inner braid, so I
use GLASS thread inside the beading as it is a good dielectric, too. Why is the
glass there? A solid TEFLON® bead can’t be processes at this size and keep
consistent dimensional linearity. The glass is the true STRENGTH member in the
beading, not the plastic. The plastic is to set and hold the shape. The glass
lets me process the beading at production speeds.
TEFLON®, really? OK, I’ll tell you. It has the lowest dielectric constant of
any solid plastic. It is TOUGH in thin walls for end product dynamic stability;
the bead should STAY round under side-wall pressure. This is a SMALL bead, so I
need that toughness. TEFLON® has high T and E’s (tensile and elongation)
properties for process toughness. We don’t have much process room, as I’ve
calculated backwards how big this bead would need to be in this design and wire
big should the conductor be based on a tube ID? There is ONLY one optimum
asymptotic wire size driven MAX AIR volume (%) based on the tube ID. The ratio
of the tube ID with the 80% air void to the inner braid surface will determine
the capacitance. Maximizing the air content will improve the efficiency of the
dielectric so the smallest loop area for inductance will also yield the
smallest measured capacitance.
is what happens when we CHANGE the wire size;
Tube ID (IN)
Wire Size (IN)
the wire gets bigger or smaller inside a given tube ID, it crowds out the air.
We COULD go drastically big in the ID of the tube and wire size (0.150” tube
ID)…but we want to hold INDUCTANCE and signal coherence in check. Inductance is
the loop area between the wire and the inner braid, and that needs to be infinitely
close, the opposite of capacitance. For a given tube ID size we want the
maximum amount of air void and the smallest possible wire to braid distance.
This means the conductor wire size has to be as small as you can process, and
with the desired capacitance. As the tube ID gets larger, cap will drop but
inductance will rise, and the opposite with a smaller tube ID. The design
target is 11.5 pF/foot on the bulk cable to assembly capacitance would be 12.5
too large a wire hurts frequency coherence so we pushed the wire size DOWN
until inductance was moving off spec relative to capacitance. A balance was
sought between wire size (coherence) and reactive variables (L and C).
can do a quick check to see how I’m doing by applying a test ground over a ten
foot sample. Using RF frequencies as a “constant” since the velocity has
stabilized to an asymptotic maximum, we measure really high VP values, ~ 87%.
This is good as it allows me to reference to end capacitance, too. I just treat
the cable like an RF cable and work the capacitance backwards from the open –
short Impedance; Z = 101670 / Cap * VP. This is about 104.6-ohms so capacitance
calculates to 11.2 pF/foot versus a measured value of 11.19 pF/foot.
know from the previous paper that Capacitance and Inductance are FLAT with
frequency, and are actually measured at 1 KHz. Our 11.19 pF/foot bulk cable
value is true at 20Hz-20KHz. Inductance is a low 0.15 uH/foot through the audio
band as well.
Capacitance @ 1 MHz per ELP 423, Agilent
E4980A Precision LCR Meter, Belden’s Cap/Ind Test Fixture
Spec for Cap @ 1 MHz: 12.5 +/- 1 pF/ft
PDB1610 B24 Cap @ 1 MHz: 11.1947 pF/ft
Characteristic Impedance per MIL-DTL-17H (ELP 142) using the included
Char. Imp per ELP 142: Imp = 101670/(C +VP)
Spec for Impedance: 100 +/- 5 Ohms
PDB1610 B24 Impedance: 104.631 Ohms
SEMI-SOLID PDB1610 finished RCA “assembly”
pf/foot IND 0.1450 μH/foot
isn’t as critical in high impedance leads as current, which is ride time
limited by inductive reactance, which is near ZERO, but in my listening test,
cable with near zero on BOTH L and C attributes sounded best, and a BALANCE
needs to be considered. The cable isn’t big or small; it is what it needs to be
to WORK. The wire size we start with sets this all into motion.
FEP tube is critical to get right. Special processes are used to keep it
on-sized and ROUND over the beaded center wire.
Shield material and design considerations.
have a core tube and know the electricals, so now what? The braid is much more
important than people think, and for a different reason than people think. No,
it isn’t shielding, either. True, a double 90%+ braid have 90 dB RF shield
properties but, I sure hope your equipment isn’t THAT sensitive to RF. Foils
are much better and more economical for RF than a single 80% braid and the
shield reaches the 90 dB mark far more cheaply.
cables are “shielded” to RF noise and IMMUNE to low frequency nose (outside their
pass band) because the shields have a low resistance to RF, measured as
transfer impedance. This is sort of like low DCR at audio frequencies, but
relates to how high frequencies work. Audio cables are not RF cables!
need to look at how unbalanced circuits work. They SHARE a ground…or do they?
They are SUPPOSED to SHARE a ground. They don’t. RCA unbalanced cables use the
CHASSIS as a ground to the wall outlet or it is floating in some cases but the
REFERENCE between the grounds is still there. In ALL cases, there is that pesky
WIRE thing called the SHIELD between the ground points on every piece of RCA
equipment you use. That wire has RESISTANCE and that resistance creates a
ground potential difference so current starts to flow between the two end grounds.
E=I*R, remember that? A VOLTAGE is impressed against the center wire and the
magnitude of that voltage is the current times the resistance. We can CONTROL
the “R” by using TWO 98% copper braids. This is $$$ to do, but it is the RIGHT
thing to do.
those braids won’t shield MAGNETIC interference. The HUM you hear is more than
likely ground loop current through the braids resistance called SIN; Shield
Induced Noise. The lower the braid DCR is the better the SIN rejection. You
need low permeability shield to block low frequency magnetic waves (anything
below about 1 MHz starts to have a considerable B-field bent over E-field).
Good audio RCA cables ARE NOT going to shield B-fields. They will shield
E-fields and reduce SIN noise.
shield magnetic B-fields a MAGNET needs to be able to STICK to the shield. This
is an indicator that the material is “influencing” the magnetic field flux
lines INTO the metal and OUT OF the air. We can manage the SIN noise with a
good ground, but true extraneous magnetic noise is still tough with unbalanced
cables. Now you know why. It’s the ground system it uses.
Jacket design and material
uses an FEP jacket for some good reasons. FEP is the most chemically inert
material there is, protecting your cables from chemicals and UV exposure
through those nice picture windows in your house. Lesser plastic material isn’t
as stable, or inherently flame retardant. Nor can many materials be used in
migration out of the cable, especially near heat, is a real issue in contact
with polyester or nylon carpet that would love to be the same color as your
cable laying on it! My previous cables were.
FEP does not have this issue and will look nice for decades to come.
Yes, it costs some more but these cables are an investment into the future and
can follow your system several steps above where you may be now. Based on
durability, stability and inertness to solvents, FEP is the best choice for the
RCA SUMMARY – Knowing that RCA cables
aren’t as “shielded” at audio as we think, what can we do about that? If you
don’t have the problem, you’re good to go! RCA is a great sounding cable by
fundamental electromagnetic design. This is why it was created. It does have
magnetic noise immunity issues, though. There is no magic to good cables; it is
adherence to strict design rules that also encompass those “magic” tertiary
variables called wire science. The same
design adjusted for a new material’s skin depth properties can be made to the
same “ratio” and match the electricals with differing wire. The layers of the onion and their thickness
can be altered (L and C values) depending on what is most audible. Tests won’t
tell you that, this comes from design experience. This does NOT mean that either L or C can be
thrown to the wind. Both L and C cause
TIME based distortions and neither is welcome in good cable.
there is the next cable I’m going to talk about that does exactly that, except
it is far, far harder to make as good as an RCA electrically. It is called the
you have spent plenty on cables you may well wonder WHY these cables are
physically as they are. If care is taken to adhere to fundamentals, there are
very good reasons for a physical design in audio cable, of both high
(interconnect) and low (speaker) input impedance types. If we look at all the
fundamental electricals through the audio band, is it any wonder every cable
doesn’t sound different? Let’s see why that might be, and no magic need apply
throughout this analysis.
is happening in audio frequency ranges?
What exactly are we “moving”
with zero distortion?
Current and Phase
Electromagnetic wave propagation differences with respect to frequency.
Impedance and matching to a load at audio.
Capacitance and Inductance with respect to frequency.
6.0 Cable Capacitive and Inductive reactance properties rise and decay time distortions.
Current normalization / skin effect.
AC resistance changes and frequency.
Cable symmetry issues.
Attenuation at audio.
12.0 Passive low pass filter effects.
we look at pure tones; sinewaves, square waves, frequency and TIME are
interchangeable. Math says that this is so, and there isn’t anything new that
explains that away. When we add TIME based distortion to the sound delivery
system our ears are quick to “hear” the deterioration in fidelity based on
frequency arrival time and phase coherence more than amplitude limitations
(attenuation). How much is a cable responsible
for this? The superposition of the 12
listed distortions (and there are more) are much more significant than any one
taken on its own. There is truth to the concept that slew rates, or how fast a
system responds (wider bandwidths), affect performance. A square wave is but a
multiplicity of sine waves. Mathematically every frequency’s characteristics,
at every point in a cable can be predicted.
Cable is far from perfect at moving electromagnetic wave through the
audio band, however well we can calculate the accumulating TIME based
distortion as the electromagnetic wave moves down the cable. Better designs
minimize those distortions and place more or less emphasis on each one
depending on the designer engineer’s concept of audible influences. The fact
remains, cable design is still driven by the DESIGN needed to reach the R, L
and C values with minimal influences on tertiary elements. Can you hear a more
fully optimized design? This is why we present these designs for audition.
1.0 ELECTROMAGNETIC WAVE
The issue –
What do we actually LISTEN to on a cable? What is the “root” reason to be for a
exist to move the “signal” from one place to another, but few really consider
WHAT that signal is. The signal we “use” is the electromagnetic wave moving
down the cable at the group velocity of propagation of the dielectric. OK, what
did I just say? Imagine our wire surrounded by a donut with a hole in the
middle! The electromagnetic wave is this donut. There is an ELECTRIC (E-field) around
our wire too, but this field is attached to the donut radially, and ninety
degrees orthogonally to the donut’s circumference. To make the E-field, take a
bunch of tooth picks and stick them all around the outside of the donut, that’s
Now we have two imaginary waves, one low frequency and one high, sitting there. To MOVE that field, electron flow starts it happening. To keep it simple let’s distort our wire to be a TUBE full of marbles (electrons) that has an inside diameter the same as the marble’s diameter. To make the magnetic field move, and drag along the E-field with it, we apply an electromotive force (electrons / marbles) to the tube. When a marble is inserted into the end of the tube, the marble at the opposite end pops out as fast as the marble can be inserted into the send end of the tube. This “speed” is determined by the velocity of propagation of the dielectric, or the tube in our case. Something funny happens with the magnetic field though; it follows the PROGRESSION of the electron (marble) flow. When the marble is half way into the send end of the tube, our donut with all our toothpicks (the B and E fields) is halfway down the cable already! When the marble is inserted all the way in at the send end, the B and E fields are at the END of the cable. So the “signal” we use travels at the VP (velocity of propagation) of the cable, and NOT the speed of the electrons at all. Those move very slowly compared to the electromagnetic B and E fields. Now we have the donut at the end of the cable. But, we won’t ever see a baker’s dozen, or zillions more moving electrons appear at the same time at the opposite end of the cable if we carry more than one frequency concurrently since every frequency has a different VP through the audio band. All individual frequencies will have significant arrival time “distortion” between frequencies. In other words, every marble that represents a frequency in my example is inserted at a different speed (Velocity of Propagation) depending on the frequency the marble represents. Ideal cable should move a signal (now we know it is the B and E fields) down a wire at the same speed and shape at all frequencies. It doesn’t.
2.0 Voltage and
The issue – Current and voltage are locked into a phase shifted
The reactive properties of inductance and capacitance are
responsible for a ninety degree time based shift in all electronics, not just
cable. There is a common ditty about the current to voltage phase relationship
that goes like this; “ELI the ICE man”. It is a memory tool to remember that
voltage (E) leads current (I) in an inductor (L) and that current (I) leads
voltage (V) in a capacitor (C).
Why is this? A capacitor has to charge with applied current
to reach a steady state voltage, so as the voltage potential increases the
current drops. The current has to be there BEFORE the voltage potential hence
current leads voltage in a capacitor.
An inductor resists current change when voltage is applied.
Current reaches a steady state over TIME with applied voltage, so as the
current potential increases the voltage drops. The voltage has to be there
BEFORE the current potential hence voltage leads current in an inductor.
These two locked-in relationships lead to all sorts of other
TIME based issues in cable and circuits. They are the variables that constitute
PHASE in an impedance trace, for instance, and reactive TIME CONSTANTS that
we’ll cover later in the paper.
3.0 VELOCITY OF
The issue – VP varies the arrival time of signals moving down a cable.
Signals should ideally leave and arrive at the same time and shape as they are
sent at all frequencies.
Audio is in an electromagnetic transition band. This is the
elephant in the room. It prevents cable from EVER being perfectly accurate when
moving low frequency electromagnetic waves. The propagation constant, the speed
at which the electromagnetic wave / signal moves down the wire’s outer
circumference, and not IN the wire, is determined by the dielectric material
that the electromagnetic wave is predominantly traveling through. We can
measure this effect directly and indirectly.
At RF, where life is way more consistent for cables, we can
calculate the velocity from the DELAY equation. For Ethernet cables the following
equation is used;
Delay EQUATION at RF
The delay equation uses FREQUENCY. This is a TIME based value so it tells us that we have arrival time issues as the frequency changes, and less so at RF, and WAY more so at audio frequencies. The table illustrates the slow erosion of speed as we reduce the RF frequency. A little change is evident but audio frequencies see much more change.
Delay values measured at RF (MHz)
Actual data shows what audio cables do; the impedance RISES as we go LOWER in
frequency, by a lot. This is because the DELAY / VP factor drops, and adds
TIMING issues to signal delivery.
Above are actual traces of how ICONOCLAST performs across
the audio frequency band vs. typical zip cord speaker wire (1313A) and out to RF,
to prove a point. The impedance increases considerably below the RF frequency
reference values. Those 87% and 90% VP factors we love to “hear”, high VP, are
clearly not valid in the audio band.
How significant is the VP change? In the example above we
drop from ~110,000,000 m/Sec @ 20 KHz to ~5,000,000 m/sec @ 20 Hz or a factor
of 22 times slower through the audio band.
To make matters worse, it is a LOG function so it is not
linear. This is what physics has thrown into the design process. Can we hear this change? Attenuation at audio
is a passive linear variable and considered to be insignificant (keep your
cables short) but every variable keeps adding up to the overall actual
Notice that the cable’s impedance, made for audio not RF, flattens
out @ ~ 50 ohms above 100,000 Hz (see the table below for the actual values).
Just because something has an “impedance” (real and reactive L and C component)
does not mean it is a transmission line.
Look at the low-frequency range. Isn’t cable supposed to be
the same at all frequencies or the same TIME base? The velocity constant at a
frequency is TIME, so the fact that we see a difference indicates a non-
linearity across the usable audio band. The
problem is that thing called propagation velocity (VP) or the speed that
information travels at differing frequencies in the cable.
The equation at audio compared to RF is more complex
(wouldn’t you know it!);
Z = sqrt((R+j*2*pi*f*L)/(G+j*2*pi*f*C))
resistance (R )
conductance (G )
Using the general simplified RF equation, where all the
extra stuff in the complicated impedance equation at audio goes to a one or a
zero and drops out, we are left with; 101670 / (Capacitance x Velocity) =
Impedance. At RF for ICONOCLAST speaker cable;
101670/(VP*45pF/ft) = 50 ohms @ RF
Solving for velocity of propagation we see it is no higher
than 45% at RF. This isn’t RF cable, and the design changes necessary for audio
are what ICONOCLAST is after. We need to
ideally FLATTEN the VP curve for audio cables to better time align the signal
in the frequency range where we use it.
The calculated graphs using a 75-ohm coaxial cable below
show that VP change as we go lower and lower in frequency. Look at the
IMPEDANCE at audio frequencies shoot way up, and the VP drop like a rock in a
pond. Notice, too, that VP begins to flatten out at 100,000 Hz, just like the
charts above on ICONOCLAST. This is real stuff, and it won’t go away…you have
to MANAGE it to a balance in each cable.
What does our measured data
show that corresponds to the theoretical chart above? Below we see several
BELDEN products measured VP drop considerably from RF to, and through, the
audio band. And, the measured values are near the exact same values I will
calculate from measurement on ICONOCLAST; ~ 5% VP to 50% VP between 20Hz to 20
WHAT CABLE Velocity REALLY DOES THROUGH
THE SWEPT FREQUENCY
The impedance goes up as we go lower in frequency because
the velocity keeps going down, and the alternative variable, capacitance, just
sits there (we’ll get to that soon). We have a differential in signal velocity
across the audio band. Also notice that typical 1313A ZIP cord behaves much
worse than ICONOCLAST™, rising to double the ICONOCLAST reference impedance
value. Be warned, audio cable does NOT respond to impedance matching like
Speaker cables are theoretically designed to be much lower
impedance, and terminate into reactive 2-16 ohm loads, and some point way north
of 16 ohms. Interconnect cable is
terminated into “high” impedance resistive loads of 47K to 120K or higher, and
should be much higher theoretical impedance than speaker cable, and the graphs
above show exactly that.
It is good to see impedance matching to a load, but other
variables are in play, and impedance matching isn’t meaningful or practical at
these frequencies and impedances. Good designs usually address ALL parameters,
Interconnect and speaker cables, with VERY low audio range
VP values show a much faster VP in the RF band. The values of 87% VP @ RF are
NOT really correct for WHERE the cable is used, but “sounds” exciting.
What do we see at RF on an ICONOCLAST interconnect cable? We can calculate what we measured in the graphs above. We can use a grossly simplified equation to predict the VP based on capacitance measurements;
101670/(11.0 pF/ft * VP) = 105 ohms @ RF
Solving for VP we get a value of 88%, using the measured values of 1 KHz referenced capacitance. This VP factor will DROP considerably in the audio range to much LESS than that. Imaginary values (L and C) stay the same from 1KHz to RF frequencies so VP is changing;
If we take the VP reduction factor of a coaxial cable into
the audio band @ 22 X lower, we see; 87% / 22 = 3.9% @ 100 Hz. Close to the
same answer in our rough calculation.
The data shows a 13X to 20X or so DECREASE in cable speed as
we drop in frequency. Signal arrival times are NOT staying in perfect symmetry
relative to the input start point. The AMPLITUDE may be near the same, but the
TIMING is certainly not. Arguments persist as to how long the cable needs to be
to her the arrival time coherence.
4.0 IMPEDANCE AT
The issue – All cables should terminate into their characteristic
impedance (not really true at audio). At audio, the cable isn’t a fixed
impedance, or even really an “impedance”. Interconnects see a resistive
“infinite” load, but not speaker cables, which see a highly reactive low
Impedance is a REACTIVE vector value. This is a dead
giveaway that we’ll have to deal with Dv/Dt stuff. All cables are a wire that
is in series with an inductor and a capacitor to ground. All three R, L and C,
keep getting bigger the longer the cable on a bulk value basis. The impedance
is a VECTOR sum of the REAL part and the IMAGINARY part. The PHASE is created
by the imaginary part of the impedance vector value. The impedance values
aren’t the same for all frequencies (see the 1 KHz and 1000 KHz chart below)
since VP keeps changing, and this is a component of the impedance value. Since
the impedance is a vector sum magnitude ratio, it stays constant for each frequency
point no matter how long the cable is. R, L and C increase proportionally.
Reactive Change with Frequency
Most of us kind of know that we are supposed to match the
impedance to the load for the best transfer of energy. We are actually only terminating
the resistive component we call “impedance” to the load; a resistor in the case
of interconnects, or a speaker load for low-impedance speaker cables. There is a reactive component that is also at
issue for good signal transfer. That reactive (usually capacitive) part of the
Impedance vector magnitude diminishes the transfer of energy in time. Audio is
not RF, so this matched resistor to resistor ideal isn’t exactly correct
anymore, even for high impedance interconnects. The physics of the velocity of
propagation make impedance matching impossible at audio as does the wavelength,
which is far, far too long to react like a true “impedance” vector.
For transmission line effects to be a factor, the cable
length also has to be at least 10X or more the quarter wave length of the
frequency of interest. This relates to the fact that a voltage change has to
happen BEFORE it gets to the end of the cable and audio speaker cables transit
times are too fast, even @ 50% VP, for this to happen.
A cable can have impedance (real and imaginary values), but
it is largely irrelevant to true load matching. There can be a signal
reflection based on the CUT length of the cable relative to the speaker. This
simple reflection can be absorbed with a ZOBEL network across the speaker
terminals if it induces amplifier oscillations. But, low cap cables are benign
to amplifiers, even with this simple length defined reflection. The cable will sound the same with or without
the network as the parallel circuit is not in the signal path. The tertiary
effect of better amplifier stability is what improves the sound with too high
At RF, a signal is “used” efficiently only when two like
resistive loads see each other. RF cables are designed so that the cable impedance
matches the restive termination load. Audio cables don’t work like this at such
low frequencies since we can never transmission-line “impedance” match to a
load with short passive cables. But, the “work” done across the load STILL has
to be resistive. The imaginary components of a vector (Impedance is a vector
sum of the real and imaginary components) store and release energy since they
are composed of reactive variables; Capacitance and Inductance, both variables,
are store and release variables of voltage and current respectively. Short
cables still have reactance.
We can see what happens at RF. The graph below shows actual
cable data of what is called Return Loss. The return loss, RL, represents the
“reflected” signal that does not transfer to the load for an RF Ethernet cable.
RL= the imaginary part that can’t do work till it is “real” or resistive.
Notice that we see several RL values “dead nuts” on 100-ohms from a low of -55
dB to a high of ~ -22dB. WHY are the RL variables not all the same? The impedance
shows 100-ohms for all those RL values. The impedance at every frequency has a
different reactance due to a lot of things too complicated to explain today.
Simply put, at the frequencies with the lowest imaginary component, more energy
is transferred to the load. In our example, if the impedance is above or below
100 ohm, and more or less reactive, the RL is decidedly worse. This is the
cause of the FAN shaped graph that we see below.
Audio cables aren’t used at RF, though, and suffer from
simple reflections more than load matching ones. This isn’t bad thing, as the
critical attributes at RF aren’t restricting what we need to do in the audio
band for better signal quality. We don’t need to worry about minute wire
diameter fluctuations that cause the above graphed RL reflections. Audio
wavelengths are too long to see the diameter variation issues so designers can
work with geometries that may not be ideal at RF, but are far more useful for
coherence adjustments in the audio band. Those adjustments still have to be
real, of course, and measured or calculated with accepted standards.
Audio speaker cable with AC signals is terminated into a
load that is resistive and reactive. Alternating current reacts to the imaginary
circuit cable values and regulates how fast, and when, we can get work out of
the cable. Some early cables were so reactive that amplifiers would shut off
using them. Even though our cable is not a true impedance we do have reactive
Interconnects see an “infinite” ideal resistive load;
47K-ohm on up, and speaker cables see a very low, and varying, reactive input
impedance (the impedance of all loudspeakers changes with frequency).
Speaker cables are
CURRENT signal devices that are designed to transfer power to an
electromechanical motor. And, a motor that constantly “changes its spots” at
every frequency as does the cable. The “argument” between the speaker EMF and
cable is complex.
are VOLTAGE signal devices terminating into a HIGH impedance resistor. We want
to transfer the signal shape and amplitude to a load. To avoid distortion(s) we
don’t want the cable or the load to mess with the transmit circuit, but they
Audio cables are way too short to be transmission lines,
needing at least 10X the wavelength inside the dielectric to be a true
transmission line. Even 20 KHz is way too long a wave length to match that
definition. We DO have simple reflections off the LOAD (speaker itself) that
cable can’t manage as the load varies with frequency. This is very different
than RF where I can make a cable nearly look like the load, minimizing
reflections. I said “nearly” as all cables exhibit reactance, a TIME based
storage of energy. Audio cables have significant measured time based
propagation error due to VP and now we add-in a rise time error from reactance.
The reactance of cable can be used to calculate “time constants”. At audio
every frequency is associated with a different constant value. We’ll look at
time constants later.
Zobel networks have been used to good effect to dampen the
cable to speaker load variation, but they are estimations of where the two are most aggressively
reflective. A Zobel network is a passive means to connect two differing but
fixed characteristic impedance lines with a resistive value. Neither the cable
nor the speaker are linear loads making it an approximation as to where to tune
the Zobel network.
Compared to our “typical” Belden cable (blue trace),
ICONOCLAST is flatter (orange trace) in velocity change as we go lower in the
theoretical impedance. This is more the result of a higher, but still low,
capacitance between the two designs. Lower inductance was preferred over
The table data below is REAL and represent what even really
good cables do through the audio band. The physics of the propagation delay
match the measurements.
WHAT CABLE IMPEDANCE REALLY DOES
The interconnect tables follow and yes, they too show time
5.0 CAPACITANCE AND
The issue – What do the reactive
variables do with respect to frequency?
Capacitance and inductance are essentially FLAT with
frequency. Yep, capacitance and inductance are, interestingly, the same from
near DC to the “sky is near the limit” frequencies. Capacitance is set by the dielectric,
assuming it is a linear dielectric material, and some aren’t (PVC).
Measurements show that stable dielectrics offer frequency linear capacitance.
Inductance is set by the distance between the wires and the loop area; it isn’t
changed by the dielectric at all. These two values are always steady Eddies,
but their time based effects on current and voltage change with frequency.
Here is ICONOCLAST speaker cable that shows L and C
and it is FLAT with frequency using
Teflon® as the dielectric.
The choice of what plastic to use sets the dielectric
constant. You want stability with respect to frequency. Teflon® has the lowest
dielectric of any SOLID plastic and thus the lowest capacitance with the
thinnest walls of any material and, it is durable. It costs a LOT to buy and
process, too. Cost isn’t why we use it,
Plastics aren’t magic for capacitance, that is just the way
it is. You want to pick the lowest dielectric constant value not just for low
capacitance, but to help offset the change in the dielectric constant with
respect to frequency. PVC dielectrics are far worse in linearity with respect
to frequency, and the slope is not the same everywhere. The chart and graph below assumes a set wall
thickness and changes to the dielectric material alone. We can alter the WALL
thickness based on the dielectric constant to get a given capacitance between
two wires. Double the dielectric constant means doubling the wall for the same
capacitance. Use the cheap stuff then? Sure, but more wall thickness increases
loop area (space between the wires) which increases inductance! Oops, we’re not
going to get zero cable reactance that way! A wire in a vacuum inside a braid
ground would be the smallest size with lowest capacitance you can realistically
see. This design would also have the lowest inductance since the loop area
would be at a minimum with the vacuum acting as a low dielectric material.
We can calculate the effects of the dielectric and
capacitance using a shorthand RF formula 101670 / C *V. We fixed the reactive
impedance to a set value, so for a fixed wall of insulation, the capacitance
rises as the dielectric constant is higher.
Since we know that the capacitance value is flat with frequency, this
applies to the audio band as well. Better dielectrics for a given wall mean
lower capacitance. This has nothing to do with Inductance, which is related to
the magnetic field lines. Inductance is related to the distance between
conductive surfaces, the less the better and field cancellation…if any.
Impedance = 100 ohms
Velocity = 1 / SQRT (E)
Capacitance = 101670 / (impedance * VP)
6.0 INDUCTIVE AND
CAPACITIVE REACTANCE VARIABLES, XL AND XC.
The issue – all cables store and release energy (current or voltage)
reactively to the frequency being electromagnetically moved through the wire,
adding time based distortion.
Look at the Impedance / Phase trace shown above on Part 3,
Velocity of Propagation Issues. Notice that the PHASE on BOTH cables changes.
PHASE include reactive components that TIME shift the signal’s ability of a
signal to become resistive (If the phase trace hits “0” the circuit is
resistive and has no reactive component).
A capacitor looks like an OPEN to DC or to very low
frequency AC voltage changes. The cable is very “reactive” to voltage changes
at lower frequencies. As you go up in frequency the cable’s bulk capacitance
looks more and more like a SHORT circuit. The cable becomes more “resistive”
looking with less reactance to voltage change. The trace explains why we can
use 75-ohms and 100-ohm loads for RF cables, they look “mostly” resistive at
Using a high impedance probe to measure the cable’s
reactance produces the traces that you see. There is little current flow into
the cable and, this is essentially how interconnects are used. The terminating
load is VERY high impedance limiting current flow. When you put a voltage
across a capacitor (our cable) it sends a momentary inrush of current to try to
fill the capacitor. Output devices loading the circuit are ideally super low
impedance to allow for this “inrush current”. Cables with lower capacitance
mitigate the inrush current issue. Current LEADS voltage in a capacitor so
there is a TIME shift caused by the cable.
Speaker cables are differing in that we don’t measure them
like they are used; terminated into what is essentially a short circuit, the
speaker. The large current flow in speaker cables responds to reductions in
INDUCTANCE. Inductors resist current flow changes and that’s what speaker
cables are trying to “move”. Voltage leads current in an inductive circuit and
again, we see a TIME shift caused by cable but the opposite reactive variable,
inductance verses capacitance, than the interconnect cable.
Also consider that in speaker cables, the most reactive
region is exactly where speaker’s impedance is also the most reactive, too. We
want is a cable that is purely resistive but that’s impossible since a cable is
a vector of capacitance and inductance.
Can we look at this another way? Yes, we can. If we examine
the capacitive reactance equations below, and stick in the values at DC (F=0)
and infinity frequency (remove F) and see what the results are we get the same
answer; reactance is high at low frequencies and lower as you go up in
Xc= ½ * pi * F * C
XL = 2 * pi * F * L
The inductive reactance is the opposite, it looks much
smaller at DC (F=0) than at higher frequencies (F= infinity). An inductor is a
SHORT at low frequencies and an OPEN at higher frequencies. Fortunately speaker
cables are relatively lower frequency making things less severe than at RF.
Cables, and all circuits, have capacitive and inductive
reactance. Capacitive reactance resists voltage change and inductive reactance
resists current change. They are both frequency dependent.
The TIME it takes to CHANGE the signal applied against a
reactive load is measured in TIME CONSTANTS. It takes about 5 to 6 time
constants to reach steady state amplitude. Our signal is also distorted the
longer it takes to reach steady state amplitude so it may get nearly as big
(we’ll pretend attenuation isn’t an issue), but it isn’t the same SHAPE. Don’t
forget, every frequency is associated with a different time constant, and the
decay or removal of the signal is the inverse. It takes TIME for the signal to
bleed away to zero and this alters the decay signal.
As frequency changes, so do the reactive variables the determine a cable’s reactive performance.
At the very high end of the
graph below, we see simply SQRT (L/C). At the low end the simple reactance
(denominator) enter in.
7.0 SKIN EFFECT
The issue – Current magnitude
normalization at audio frequencies. Is this real?
There are several ways to calculate skin depth, and they all
will yield the same answer. Impedance / RL can be derived from several
inter-related factors and so can skin depth. It is real, and it can be managed
to control phase distortion.
We all know about skin effect, but WHAT exactly is it doing
at audio frequencies and is it real? Yes, skin effect is real at audio and all
industry accepted calculations show that it is. The definition of skin depth is
the point inside a wire where the current decreases to 37% the surface current
magnitude. Skin depth is always the same depth of penetration no matter the
wire size. Skin depth will vary based on the material’s electromagnetic
properties and the frequency of the signal. For audio we calculate @ 20 KHz.
Looking at COPPER, we
would calculate 461um (0.0181” depth).
If the skin depth at a given frequency is 10-mil on a
100-mil wire the 37% current point is well near the wire’s surface, it’s just
10-mil away in 50-mil radius. If we
halve the wire size, the current magnitude is larger through more and more of
the wire. Each time we decrease the wire size, the larger the current magnitude
becomes across the wire relative to surface current. In our 18-mil skin depth
wire example above, the current in the “center” of a 36-mil wire will see 37%
the magnitude of the surface current. Making wire smaller will INCREASE the
current magnitude in the wire’s center to be closer and closer to the surface
current in the wire at higher frequencies.
AC resistance involves FREQUENCY which is a TIME based
RAC= (RDC) (k) SQRT (Freq)
K is a wire gauge factor that involves skin depth.
Freq is in MHz.
The internal wire impedance (AC resistance) is driven by the
INTERNAL magnetic field’s relationship to inductance. Inductors RESIST
instantaneous current flow and have higher “resistance” as AC frequency goes
up. Current flows in the least resistive part of the wire as frequency goes up,
so it reaches the surface where the self-wire inductance is nearest to zero.
Once we flatten the velocity change as best we can with a
good dielectric design, we need to ALSO time align the effects of the
dielectric at ALL frequencies using SMALL wires. Small wire improves arrival
times as it forces the effects of the composite dielectric speed to be more
uniform, as best we can, at all frequencies. This counters the skin effect
problem that moves the current density magnitude to the surface of the wire as
frequency goes up. Smaller wire increases the current magnitude (arrow length)
in the wire center region to make it more efficient at time alignment.
One BIG wire
More SMALL wires
Even if we have the SAME current magnitude throughout the
wire at all frequencies (impossible unless our wire is one atom in size) the
velocity of propagation of the electromagnetic wave energy is STILL different
at every frequency! But the ears say if we MANAGE the problems, our cables can
sound much better. I took the time to measure all of this and flattened the
impedance trace as much as I could. The VP changes less with frequency the
flatter the impedance curve. Capacitance stays the same at all frequencies, so
this VP is therefore changing less the more consistent. Smaller wires are more consistent
dielectrically at all frequencies.
Bigger wires will cause even more signal speed change
relative to frequency because each electron’s is far smaller inside the wire.
Each magnetic field contribution changes velocity the closer or farther that
electron is away from the dielectric material. When a current is applied
(electrons start moving) an inner wire located high frequency current mode
travels slower than the same frequency signal on the outer wire surface and all
these current modes are superimposed one on top of the other. This is called
Not all signals at the same frequency arrive at the same
time, it depends on WHERE they traveled (MODE path) through the wire and what the velocity of propagation is from the
geometric perspective. The lower in frequency you go the less you can change
the group delay since the current density through the wire is more and more
The overall magnetic field is a summation and superposition
of ALL the moving electrons, the whole “group”. This is also why air is often
used in interconnecting cables to mitigate the dielectric’s impact on the
signal, and why you see more small wires in speaker cables. Electromagnetic
field uniformity in the dielectric is important. The overall audible
improvements are more debated. But, there is science involved in the
The issue – dielectrics can impact weak electromagnetic signals
disproportionately. Electromagnetic fields are squared law fields, and are most
influenced by dielectrics nearest the wire. Weaker electromagnetic fields are
most susceptible to dielectric distortions and the group velocity is mostly set
by the strongest signals dielectric medium.
Using too many small wires splits up the current and starts
to allow the dielectric to influence the sound more and more, negating the
“advantage” of dielectric uniformity. The electromagnetic field is strongest
nearest the wire, decreasing with the square of the distance moving out away
from the wire. The electromagnetic signal moves from being “in” the dielectrics
to being around it. The signal propagation speed is an average of ALL the
dielectrics, with the material the stronger fields reside in have the most
influence on the average of the “group”.
Four-fifths or more of the current magnitude at audio is
below 3 KHz. Some call this the spectral power density, or roughly where the
most energy is being placed. The electromagnetic energy does not STOP in the
plastic or air. It emanates out in an inverse LOG power decay THROUGH all the
materials it encounters along the way. The predominant material VP effect
occurs CLOSEST to the wire. The smaller the signal (interconnect cables) the
bigger the effect of the immediate dielectric nearest the wire.
Weaker signals will be impacted by the dielectric’s effects
more than stronger ones, as they decay to far weaker signals moving away from
the wire. The speed is more and more determined by the dielectric near the wire
as we go up in frequency. Interconnects see little of the plastic out away from
the bare wires as the field decays so quickly, but, the smaller the
electromagnetic signal, the MORE it is influenced by the material nearest the
wire. That superposition of materials SLOWS the signal (air to plastic) or
speeds it up (plastic to air) relative to just the initial material’s
We can see this in actual practice as the “group” velocity
of all the materials on Ethernet cable shoes a value SLIGHTLY higher than the
dielectric (66%) itself, and measures 71%. The signal is in the “air”, a good
dielectric” and this influences the overall signal speed.
ICONOCLAST interconnect design switches this around, and
puts the AIR nearest the wire, where the signal strength is highest. This
negates the outer plastic dielectric’s contribution to the group velocity, so
we see a higher 87% value at RF. This translates to lower capacitance number
where we use the cable in audio applications.
The VP speed variation caused by the “composite” velocity is
complicated by the fact that the LOWER in frequency you measure, Mother
Nature’s devious plan slows everything and this time shifts the audio band.
We can’t change the fully diffusion coupled (same magnitude
current through the wire) low frequencies, so we try to time align the faster
upper frequencies. At RF the upper frequencies are “on” the wire surface so the
dielectrics affect them nearly 100%. At RF this is fine because it is near the
same VP at all RF frequencies. At audio, we want to move most of the high
frequencies AWAY from the dielectric so the speed is closest to the lower
frequencies. We already know that the VP
is faster the higher in frequency we go so this messes up the signal arrival
times. The only good way to slow the
upper frequency magnetic field is to make the wire smaller so less energy is
JUST at the wire surface nearest the dielectric. More current is “in” the wire
versus “on” the wire based on skin depth.
Can we overdo field current normalization? What if we could
make a wire one atom wide? Now, the impact of the DIELECTRIC is as big as it
will ever be and with a really, really small current in each wire. The total
current will be the sum of all the wires we want to use in parallel. The more wire
you use, the smaller the current in each wire. Current is the number of
electrons past a point with respect to time. Well, we have ONE tiny electron
moving in each “wire” and THAT is as small a current as you can have! Model a
weak signal, and the electromagnetic wave is so weak it never really leaves the
dielectric, whatever material the dielectric is. The dielectric better be
really decent as it is hugely involved in capacitive rise time (calculated
capacitive reactance rise times constants) signal arrival time (velocity of
At very high frequencies, and if the wire is infinitely big,
we see ONLY the dielectric as the current is at the surface (skin effect).
Likewise if the wire is infinitely small we AGAIN see JUST the dielectric (no
skin effect can happen). Between the extremes of wire size, somewhere, we can
alter the arrival time of the upper frequencies with wire diameter and
Interconnects are much easier, but not real easy, as they
terminate into a high resistance, nearly open looking circuit. The reflections
off a CONSISTENT resistive load of 47K-120Kohm aren’t as bad as the mismatch
speaker cables experience as BOTH the cable AND the load are in constant flux.
Worse, the speakers change by design! The seemingly high measured impedance
slope of RCA or XLR interconnects in the initial graphs aren’t as bad as they
seem. Not only are the “impedances” not real at audio but you have far bigger
issues with the non-linearity of cables loading the output devices in your
preamplifier. Trying to match ideal infinite input impedance on RCA or XLR
cable would mean tiny capacitance
values. We go as LOW in capacitance as we can to allow the output devices to
see an easy load. This is way we shoot for keeping capacitance reasonably low.
Massive signals in the speaker cable are less impacted by
the “composite” dielectric speed. The electromagnetic field will travel at an
“average” of all the stuff it is moving through, so the better the “average”
material is that the electromagnetic field is in, the FASTER the signal
travels, and the less TIME the signals have to become separated as they travel
down the cable. This is the time and distance story problem.
PC’s stopped using FLAT cables because the signal arrival
TIME differential got to be too high. They went to SERIAL digital designs, and
re-clock the data from memory. This at first seems counterintuitive, adding the
re-clocking circuit, but unless the TIME can be managed, you’re screwed. Faster
is better but I’d take SLOWER and the SAME in an instant! This is the “keep
cable shorter” thing, but to be LONGER we have to be FASTER, too, if time
errors are to be kept low. Mother Nature says we get a raw deal in the audio
band verses RF.
In speaker cable, the stronger low frequency electromagnetic
waves emanate into the air through the plastic dielectric more than the higher
frequency signals so they are theoretically aided by the air around the wire
(superimposed dielectric value) more than the weaker high frequencies that see
more of the slower plastic dielectric. But the VP erosion as we drop in
frequency eats-up that advantage in the low-end. It’s there, but small. The
problem is that the low frequencies still drop in speed way more than the air’s
addition to the overall speed. Seeing more air as we go lower in frequency
speeds the signal up relative to the faster high frequencies and offsets some
of the problem…but it never aligns it away to zero. The VP still marches slower
and slower as we go lower and lower in infrequency. Arrival times are more
important than SPEED down the wire.
The highest frequency carried in the speaker cable is most
fragile, but compared to interconnect cable, it is relatively robust. The high impedance interconnect cables are
yet another problem. ALL the signals are VERY, VERY low current electromagnetic
field energy states. Here, I need the BEST material possible to time align the
energy field “whipping” (slowly whipping) down the wire; air. The VP is the
inverse of the dielectric constant so we want a fast dielectric and the lowest
associated capacitance it can also provide. This is why I HAVE to use AIR core
designs to properly time align the energy AND use SMALL wires to better
distribute the dielectric’s effects at ALL frequencies nearest to the same
composite velocity. The third leg is to decrease output device capacitive
loading. Air helps mitigates velocity variation across the frequency band that
is the bane of audio signal transmission. It incidentally also pushes UP the
“impedance” to better match the load, the opposite of a speaker cable. I’d be
wary of that improvement as we need to be aware that audio isn’t a transmission
It seems counterintuitive to use air, as it speeds up the
higher frequencies relative to the lower frequencies (makes the difference
worse) but the capacitive reactance influences rise time error if you let it
get too high. The propagation time and the rise time need to be balanced,
somehow. There is no perfect solution.
We call, it “sound quality” when we use the cable, but it
really is the arrival time alignment of all the signals. The human brain hears
superimposed time alignment and amplitude preservation first, everything else a
distant second. The argument is: does this make a difference?
CABLE RCA to XLR MATCHING
The issue – Changes in electromagnetic properties between interconnect
cables types can alter the ideal “tone” that was intended.
An often ignored issue is, what do you do with a really good
sounding RCA cable? Why not make a really good sounding XLR that’s the same
reactive measured design? Most RCA to XLR cables never match. ICONOCLAST is no
accident. I purposefully designed the RCA and XLR to be the exact same reactive
match and thus the same “quality” of sound through the channel. The above
impedance chart that we saw earlier shows both the RCA and XLR. Look closely,
they are electromagnetic buddies.
Does that make a difference? If you have a very good RCA
design, it sure can’t hurt to start there on the XLR!
10.0 CABLE SYMMETRY
The issue – how to make complex cable’s cross section look like one
simple wire electrically, and every wire sound the same?
Matching multiple wires into a complex structure isn’t easy
to do well. The ideal cable is one wire that is exactly the same as the
opposite polarity wire. To meet other objectives, we usually have several
More small wires will make a nice big capacitor (wires with
a dielectric between them) and trash reactive signal conversion to resistance
products. Inductance will inversely follow capacitance, messing up the current
delivery, and to get BOTH intrinsically LOW, you can’t go “whole hog” on the
opposite variable. The two variables are tied together inversely. Rats! A
suitable compromise must be reached? Yes, audio is a compromise, as we are
seeing. BOTH L and C need to be low in value and good design manages this.
Trade-offs for better sound can, and should, be logically explainable.
The less understood variable is Inductance. This variable is
a big contributor to more wires. We all think, “capacitance” for audio. Realize
that if we had NO inductance, we could separate the wires as much as we wanted
and eventually have no capacitance (outer space actually has capacitance, so
that’s impossible, too). The reduction of the magnetic fields by proper cable
geometry reduces the inductance, allowing a larger wire center-to-center
distance for low capacitance. The LOWER
the electromagnetic field, the LARGER the loop area can be (lower capacitance)
for a given inductance and vice versa. Too many cables ignore getting the
electromagnetic field as low as possible. The higher the current (speaker
cables) the more field energy you need to eliminate. Wires with low electromagnetic fields and
small loop area have the lowest inductance.
ICONOCLAST’s speaker cable design balances out the wires’ proximity to
one another so as to not “rob Peter to pay Paul”. The unique weave pattern
increases the average wire C-C (Center to Center) distance creating a wire
pattern that CANCELS the electromagnetic field while increasing the average
spacing for low capacitance. Cancelling the field energy allows me to also
lower inductance which would be impossible to do with JUST wire spacing for
capacitance alone. No magic need apply.
EVERY wire in a cable has to be the same wire if you use
superposition of the electromagnetic fields traveling down each wire. This is
why symmetrical cable designs are used to efficiently remove reactive time
alignment issues. I measured the reactive time based issues on other designs
and they all came up short. Capacitance and Inductance have to be the same on
EVERY wire to as tight a manufacturing standard as is possible. Multiple, and
differing wire sizes are too complex to align things nearly as well. The
signal’s SPEED has to be best matched at all frequencies and not just the
physical wire length. The wire’s “signal length” is the problem. Use too many
non-symmetrical, differing sized wires and this is all but near impossible to
do with all the variables involved. I call this type of mixed wire cable,
“cable in a cable”. The effect is a kindergarten lunchroom in the dark; a mess.
In passive cable, you can’t force the highs to go in the
small wire and the lows in a bigger wire, and adjust the wire lengths to offset
the VP changes. The ENTIRE spectrum goes into EVERY wire, so now we compound
the time based issues. Only active electronics can separate the spectrum, and
that’s a problem too.
Does ICONOCLAST remove the “cables in a cable” problem? Only
one way to find out and that is to MEASURE them. The data is showing each
polarity with 12 two wire BONDED pairs, 24 wires in each polarity, and 48 total
wires in each cable;
175.3815 pF (X) +175.3815 pF = 180.1803 pF
175.3815 pF (X) + 171.4507 pF = 171.4507 pF
X = +2.74 % and – 2.29% variation between wires, or, they
are ~ 97.5% the same.
I would say yes, I got it right.
For ICONOCLAST speaker cable I set my design goal at no more
than 50 pF on capacitance and 0.1 uH/foot an inductance (45 pF and 0.08 uH
typical). On the interconnect I set the goal at 12.5 pF and 0.16 uH (12.0 pF
and 0.15 uH typical). This is WITH
connectivity and tested to prove it.
The complex electromagnetic designs of the RCA, XLR and
speaker cables allow ICONOCLAST to exist. The RCA is the most pure electromagnetic
equation that I have to work with and defines the interconnect cable problem.
How do we reach the greatness that a PROPERLY designed RCA does in the XLR
design (matched impedance / phase)? How can I convert the small signal world of
the RCA and XLR into the large current world in the speaker cable (low
inductance with still low capacitance)?
11.0 ATTENUATION At AUDIO
issue – is it mostly LOG linear so we can’t hear it?
If it is true that we can’t hear LINEAR attenuation
(measured Rs values say there is non-linearity) or TIME based issues in audio
cables, WHAT are we hearing with optimized designs, i.e. those that try to get
L and C to near ZERO as we can and with low time based issues? The design goal
difference in ICONOCLAST is TIME based and I’m not so sure that the
inaudibility of difference values of 5-10 micro seconds is correct. Linear
attenuation, I agree, is MUCH harder for the ear to pick out in typical cable
lengths. I said LINEAR LOG type decay.
Rest assured, if there is snake oil in these products it
sure looks like physics to me. All the above data is measured and real. The
question remains, WHY do the cables SOUND so much better if TIME based issues
aren’t audible? WHAT are we hearing, then? The reactive TIME altering L and C
along with the VP change with respect to frequency seem to be the difference in
cables, and audibly so. Linear attenuation can’t account for the differences.
Series resistance says that that factor isn’t as linear as we’d like, either.
There is a measurable difference in cables resistance across the audio band.
Is attenuation linear? I measured the Rs (series
resistance), with respect to frequency, of ICONOCLAST and saw a significant
CHANGE in attenuation with high quality R, L and C. Look at standard 1313A
speaker, 10 AWG Zip cord style cable (red trace). ICONOCLAST flattens resistive
non-linearity artifact, and the interconnects are both flat to 20 KHz human
hearing test point. Still, look at the
UNITS; it isn’t a wall of lost energy above 20 KHz.
12.0 Low Pass Filter
The issue – Cable is a low pass filter, and rolls off the frequency at
the frequency of the filter’s cut-off; Fc.
How does this change what we hear?
I saved this one for dead last since it was even overlooked
on my categorization of issues with audio cables. You’ll see why in a moment.
Here is the basic circuit. There is actually a SMALL
inductor in series with the resistor but notice that it doesn’t appear in the
equation that defines how the filter will behave, and is omitted. There are
circuits that involve larger inductors, and unless we have a resistor to
ground, they won’t apply to “cable” filters. Well, decent cable anyway.
The capacitor is a reactive device, like I’ve mentioned
before, so its properties change with frequency as does an inductor. A
capacitor eventually looks like a short to ground (capacitive reactance value
keeps changing) at higher frequencies so the signal energy takes the path of
least resistance through the capacitor to ground. R is in Ohms when capacitance
is in picofarads (pF).
The good thing about almost ALL
audio cables is that the roll-off properties of the filter are WAY above the
audio band. Yes, a first order filter will change the PHASE at the -3dB
attenuation point by 45 degrees, and time based distortions are more audible
than the roll-off attenuation. First order filter attenuation nor phase changes
are going to be an issue, theoretically.
Typical ICONOCLAST™ R, L and C Variables
The RCA shield “goes away” as
it is such a low resistance in series with the center wire, leaving essentially
the center wire DCR.
The XLR uses TWO 25 AWG wires
in parallel for each polarity, so the resistance is HALF the two wires, or
about the same as a ~22 AWG wire.
Calculating Fc we arrive at;
15.5 GHz for the 5 foot RCA.
36.4 GHz for the 5 foot XLR.
15 GHz for the 10 foot Speaker Cable.
The real problem with cable is that it can load down the
output op-amps with too high capacitance and change the frequency response and
possibly phase response. Some really high capacitance or high inductance
speaker cables can bug the heck out of power amplifier output stages, too. But
these problems aren’t filter problems, but bulk capacitive or inductive loading
problems on the output circuits.
All circuits “push back” below their operating region into
the pass band but a rule of thumb is to keep the fc pass band 10X or
more above the circuit’s operating frequency. We surely are meeting that
requirement with any decent cable, even zip cord.
Many outside this sub-discipline of engineering will STILL
insist that electromagnetic field time management and time alignment are not
important, and that only the bulk R, L and C matter. The ear is a time domain
instrument and readily time aligns the signal to the natural world we live in.
EVERY effort was made to pay attention to TIME domain issues in audio cables
and attenuation non-linear artifact. There are a myriad of ways to lose track
of TIME, and an audio cable is not a good place to make mistakes.
Consider all the measured and factual information above on
cable design and then ask yourself why cables sound different. Why wouldn’t
they sound different given how complex
it all is? True, poorly made cables all fall into a bunch of warm and
soft sounding products. Elevate the engineering and they indeed measure
different. The above is 100% true for ALL cables, if I may add. If I
mischaracterized a topic then, of course, only my cables are affected! All the
cable designs in the ICONOCLAST line are under US patents.
I hope my cables bring years of enjoyment to you, and NEVER
a feeling of complacency in what was provided to enhance your hobby’s (mine
too!) pleasure. The search is constant to try to align TIME based issues to
arrive at the best sound possible. The bad layers of the onion can’t be
removed, but the order and thicknesses can be altered. Signal coherence is both
arrival time and amplitude time dependent. Passive cable won’t allow
perfection, just a lot of hard work to manage the ill effects that Mother Nature
threw our way.
“Sound Design Creates Sound Performance”, and this means
driving down all measurable variables to the lowest possible balance we can
achieve. Does this make better sounding
Galen Gareis Principal Product Engineer ICONOCLAST Design Engineer
We tend to believe
that SHIELDED cables are superior to UNSHIELDED cables but the opposite is true
from a signal transmission evaluation. Why we feel SHIELDED is better is
because we overestimate NOISE ingress (outside the cable into the cable)
Cable electrical is determined by the primarily REACTIVE
variables that change signal shape arrival times. SHIELDING is to be considered
a necessity if, and only if, the ingress noise is more damaging than the time
based errors and physical size shielding imposes on cables. Why even have
shields if it doesn’t HELP improve the signal integrity from one end of the
cable to the other?
Capacitance is derived by the relationship of the shield to
the signal conductors in cable. The shield is usually at GROUND potential to be
a low impedance path for noise, so far so good. The bad news is that the CLOSER
a shield is to the signal wires, the more the cable varies per unit length in
measured electrical values of capacitance. It isn’t the same cable all along
its length from shield geometry variation, and the variation is much more
aggressive the closer the shield is to the signal wires. Capacitance, and thus
also inductance, change with smaller physical changes in the cable.
If you want to keep cable size SMALL, a shield means much
higher CAPACITANCE. And, a smaller size WITH that higher capacitance means a
larger per unit length variation in measured electrical. Even with AIR as a dielectric, we will see
much higher capacitance, and have a harder time controlling it with shields, so
we better need one for the function of the cable, and where it is used.
The following calculated table shows that the DIELECTRIC
in-between the shield and the signal wire can REDUCE capacitance, but only to a
point. It cannot remove the shield to conductor physical variation, which is
built into the DESIGN, good or bad.
How bad is the actual variation between the shield and
signal wire? This exact question was discussed when ultra high-speed
communications cables were being developed. Do we control the center-to-center
spacing in a BONDED PAIR over all else, or do we control the shield spacing and
geometry AROUND that bonded pair? BOTH will influence the final impedance, and
its variation. Which is really the bigger problem? Can we make better cables
managing what really makes the biggest difference, and reserve the less
aggressive physical attribute for higher performance requirements? This can
make the AVERAGE level of performance much higher at a much lower cost than
blindly trying to manage every variable all the time without a firm reference
to the cable’s final electrical values and variations.
Here is that exact analysis;
To demonstrate the
effectiveness of conductor center to center (C-C) in an ISTP cable, the example
below shows a change of C-C from 0.055” to 0.072”, holding a constant 0.061”
insulation diameter. This simulates a conductor with poor concentricity within
a well-controlled and constant insulation diameter. The impedance is nominally
102 ohms with a 0.061” C-C spacing and changes ever so slightly as the
conductors are spaced closer, or farther apart.
The shield inside dimension is a constant 0.122”. Under these
circumstances, the impedance goes from just over 101 ohms to just over 97 ohms.
A total impedance spread of about 4 ohms.
The significance of
the calculations is the relative insensitivity of impedance value with changing
C-C spacing compared to the variation in diameter of the shield, both of which
affect impedance variation with frequency. The impedance versus shield spacing
graph shows how severe the impedance change is with ISTP shield inside diameter
(I.D.) changes. Just a 20 mil change (0.120”-0.140”) moves the impedance almost
14 ohms. Our specifications allow only a 15-ohm swing.
The control of the effective shield diameter is three and
one-half times more sensitive than the C-C spacing of the conductors in ISTP
cables. Or, shield tape control is much more important than insulation
centering or backtwisting to compensate for off-center conductors. Also notice
that the closer the conductors move towards the shield in the IMPEDANCE VS
CONDUCTOR SPACING chart, the more Zo changes. When the conductors are 0.055” to
0.065” C-C, the impedance varies by less than one ohm. In contrast, when the
conductors are near the shield in the 0.065” to 0.072” C-C range, the impedance
changes 4 ohms. Unless your C-C is well out of spec (we have a 0.01” variation
with little change in impedance in this example) good shield dimensions are
much more important.
In contrast to ISTP cables, the UTP cable example shows
how profound the impedance impact is when the C-C changes just 11 mils compared
to 17 mils in the ISTP example above.
Where the ISTP cable had about a 4-ohm swing, the UTP cable has a 60-ohm
swing! In UTP cable, ground plane consistency is inherently stable because it’s
the metallic area around the cable which, under normal circumstances, is
perceived to be infinitely far away by the cable, too far to effect the
electrical to any significant degree. So the crucial variable in UTP cable for
consistent impedance is the strict control of C-C. This is why Belden’s
patented bonded pair technology is so important in UTP cable designs.
Impedance is, after all, a function of the Inductance,
capacitance and dielectric values. The impedance variation, and even at each
frequency in the audio band, changes with the dielectric and the spacing.
A cable with NO SHIELD, sometimes called UTP, does have a
reference ground “around” the cable, the environment. But, the capacitive and /
or inductive coupling are so far away that changes in the “reference” are
Shields have to pull their weight in signal integrity
improvements compared to cables used without a shield. If we have no external
noise, SHIELDS ARE WORSE than no shields! The math of cable electrical
stability firmly squares that up, per the data shown above.
This forces the consideration of NOISE. It even considers
HOW noise is transferred into (ingress) a cable, and even if the cable itself
is the source of NOISE for other external devices (EGRESS).
First, let’s be super straightforward about this from a
25,000-foot view. The closer a shield is, the capacitance value is high, and it
varies the most around the average value. Knowing that the proximity a shield
has around the signal wire can really upset the cable’s uniformity of
electrical, and how uniform we can engineer them, would we not want to use
designs that NATURALLY calculate an advantage to use with shield? Yes, we
To keep this easy, look at coaxial cables. This technology
HAS TO HAVE a shield to work. A signal wire surrounded by a shield. The signal
waveform travels along the wire surface, and under the shield surface and
inside the dielectric as a TEM (Transverse Electromagnetic Wave) wave. The more
perfectly round the inner surface of the shield and the outer surface of the
signal wire, the lower the capacitive and inductive variation and thus a lower
For signal transmission, we use 75-ohm cable (77-ohm is the ideal) and for
power 50-ohm Cables (30-ohm is the
ideal). Approximately 53.5-ohm military RG cables came about because it is the
mean between 33 and 77. If we freeze the materials we use to make the cable
(same plastics and metals) we will see that a 75-ohm cable has a larger
dielectric layer (lower capacitance) than the 50-ohm cable.
This is nice, since the farther away the shield is, the less
a given VARIATION of the shield changes the electrical stability. Reactive
variation impacts small voltage signals far more than larger 50-ohm power cable
applications with much more robust signal levels.
In a 50-ohm power type cable, we have a shield that is far
closer to the signal wire. This seems like a problem and it is, but the SIZE of
the signal is vastly larger than the NOISE. We can overcome the noise with a
larger signal, and even the return loss caused by more variable impedance can
also be mitigated with the size of the signal on power type coaxial cable.
This is simply the signal to the noise reference working in
our advantage in each design.
Voltage signal cables, dB or dBm, need shields farther
away (higher impedance) and it so happens this is the case with 75-ohm cables,
reducing capacitive coupling of noise.
Power signal cables, often in WATTS, need closer
shields for energy transfer (lower impedance) but this allows more capacitive
noise coupling. 50-ohm cables use more robust signals to overcome the noise.
This is like a low impedance speaker cable’s signal WAY over the terrestrial
There is NO EXCEPTION, lower impedance cables are much more
subjective to NOISE than higher impedance cables with the same noise ingress. We must fit the signal levels to the
impedance for ideal overall performance. 75-ohm cables are far better for
low-level signals as they capacitively couple less noise, as the DISTANCE to
the shield is larger.
To put the signal in perspective to NOISE, look at the table below.
Digital data cable go 100 meters / 328 feet with
over 23 dB of attenuation at 100 MHz and with ZERO errors due to external
noise, with UTP designs. Audio cables go mere feet, and yes we seem to want to
be the underdogs of signal integrity but we aren’t, and that’s a really good
The integrity that even a MC phono cartridge’s
0.35mV signal represents to the noise is in our favor. The robust signal even covers up POORLY made
SHIELDED cables. Do the shield really right, and it can help some RF, but
usually in a good unbalanced RCA system a RF bleed capacitor routes RF to
ground through the cap somewhere in the ground.
Coaxial cables need shields to work, and they need
shields to be super low DCR to prevent ground loop differential currents
between devices. The GROUND is shared in coaxial cables at uneven ground
reference points. RCA grounds have resistive differences. This can cause signal
bleed between channels. A BIG part of an audio coaxial cable shield is to
mitigate ground potential differences, and not to “shield” ingress.
A balanced XLR uses a SHIELD, yes, but it is NOT a part of the signal path, and
each right and left signal doesn’t share the virtual ground between the
differential voltage signals. Each amp has its isolated virtual SIGNAL ONLY
ground reference. There can be no inductive or capacitive coupling of right and
left channels. Unhook the GROUND on a XLR and it will work, with MAYBE a
slightly higher SN ratio. The outer shield simply knocks down the noise ingress
at RF, if any is there, so UNBALANCE in the pairs mitigates to a lower residual
value. One-percent unbalance of a small signal is better than one-percent of a
This is the true advantage of XLR cables over RCA.
Both have good RF noise immunity with the XLR having far superior signal
channel isolation and…low frequency noise isolation.
Since an XLR FLOATS the virtual ground independent
from any other signal, noise is the same on each leg, so it cancels. We see the
“difference” of each leg as the signal, which doesn’t change potential. This
includes magnetic and electric fields. Coaxial cables can’t shield magnetic
fields since copper is “invisible” to 60 Hz magnetic interference.
ICONOCLAST™ uses SHIELDS, but the WAY we use shields
insures geometric consistency to the signal wires. Care was taken to insure a
good BALANCE within the XLR signal wires so even if a shield is broken, little
performance impact will be measured;
Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter,
Belden 4TP Cap/Ind Test Fixture, all tests performed on a 20ft specimen.
Pr to Pr(star quad) – 10.4113 pF/ft
UnBalanced: Pr 1 to Shld – 401.9868 pF/20ft
Pr 2 to Shld – 405.9738
Cap UnBal ((diff/max) * 100) – 0.98%
Requirement – 3% maximum
SHIELD TRANSFER IMPEDANCE – This is
a measure of the cable’s shield impedance in milli-ohm/meter. The lower the
transfer impedance at a specification frequency the better the shield at that
frequency. It is frequency and design dependent. The current traveling in the
shield times the transfer impedance produces a interference voltage product to
ground in the shield, E=I*R where R is the transfer impedance.
SUMMARY – Shields have to be considered relative to noise and the
resulting S/N ratio since the application of a shield is ALWAYS a negative
variable to signal integrity that has to be over weighed by true noise
mitigation. If noise is paramount over the signal, then shields are a necessary
requirement. If shields are a part of the signal path, then the noise they
generate has to be mitigated with shield DCR.
ICONOCLAST uses shields properly,
and insures that the negative influences are geometrically stabilized, and
measured for performance in both RCA (DCR) and XLR (unbalance percentage).