Parallel Speaker Cable Wiring Analysis

Blue Jeans constantly tries to drive down the cost of our hobby, and of course we do that with our wire products. The 1310A is a stepping stone to more complex and expensive designs, called ICONOCLAST(TM).

Four separate and different cable examples demonstrate how parallel cable halves L and R but doubles C. The physics delivers that every time all the time.  If this theory is TRUE, it impacts EVERY cable put in parallel. Usage is upon the user to evaluate the change in capacitance based on the root cable electrical and length.

. This paper will cover exactly WHY we added 1310A and even better, will show you how to improve speaker cabling to get meaningfully better measured electrical.

What exactly is dual cable bi-wire anyway? Here is the full Monty application pictured below. 

In this application we see TWO cable sets parallel to the speaker bi-wire terminals. Each speaker terminal sees FOUR wires in parallel. Why would you want to do this? This report will investigate the double and single parallel cable arrangement and show a full set of measured results using four different speaker cables so we can assess the advantages. The copper draw science — TPC, OFE and SPTPC and other metallurgy — have no impact on parallel wires’ L and C, just R and that is very small. L and C change a lot as we will see.


ICONOCLAST is a woven set of two BONDED PARALLEL wires in PARALLEL with several more, 24 or 48 total parallel wires per polarity. We already use parallel wires to reach electrical properties you can’t approach with just two wires and aren’t really doing anything new. Electrical circuits see different parallel current paths with ALL cables that use insulated multi-wire arrangements. Even though it looks like ONE wire on the outside, it is several in parallel on the inside.

The series II ICONOCLAST pushed the machines as far they can, to reach proper electricals with smaller 28 AWG wire. To get that last bit of DCR well below audible, TWO cables are used in parallel on the bass side. The series I ICONOCLAST already has low 9600 CMA DCR and is full range, so to speak.  DCR in both designs improves paralleled, though. We shall see what this is all about as we work through the actual measurements in this study, so we can see for ourselves what is happening.

Some cables are high capacitance. One cable can load the amplifier far worse than low capacitance paralleled wires. The TOTAL capacitance needs to be calculated to understand the circuit behavior.  A single 1800 pF/foot cable is a higher reactive load on your amp than paralleled 1313A, 1310A or ICONOCLAST series I or II cable.  Yes, some cable designs are just long capacitors. A capacitor has low inductance and that’s the single-minded design goal. DCR is just a function of the CMA area of all your wires added up. What if we want to drive down ALL the variables such that the balance is better for us? Keeping capacitance low is important so the base designs are all low capacitance and inductance. We will see that capacitance ADDS in parallel. One cable, or more added in parallel, adds capacitance from all individual circuits (every single little wire is a R, L and C network).


How does all this get the right signal to the right place? Electricity is lazy and is frequency dependent, and will ALWAYS reach the LOWEST energy state possible, static or moving. This is how a cross-over divides up the signal based on the input impedance, or resistance, across a specific frequency range. The electricity will follow the easiest path.

When a frequency leaves the amplifier terminals it (believe it or not) looks at the far end for the LOAD at that specific frequency and takes the path of least resistance. Consider a speaker with a 200 Hz and a 2000 Hz cross-over inside. Signals that leave the amplifier below 200 Hz go to the woofer, signals between 200 Hz and 2000 Hz go to the mid-range and signal above 2000 Hz go to the tweeter. The electrons all know the easiest path to the right load based on frequency. The speaker’s internal cross-over is designed to tell the electrons where they need to go with frequency. We aren’t doing anything different than with ONE wire. The speaker still divides it all up based on frequency. What we are doing, is adding a better measuring cable to the circuit with external parallel wiring.


In the pictured case above there are TWO identical cables in parallel to each speaker section, woofer (200 Hz and down) and mid / tweeter (200 Hz and up). The cables are 4 feet long and this length has to be considered, as we shall see. The cables are all in parallel at the amplifier even though we have TWO sets of terminals. They are still connected in parallel inside the amp…it just makes it easier to hook it all up.

What happens when we parallel cable? From a textbook viewpoint the resistance halves (twice the wire), the inductance halves (half the current in each wire) and the capacitance doubles (twice the plate area). If we add TWO parallel cable sets in parallel, we further keep halving R and L and doubling C. Does cable really behave like the parallel theory says?

To test the cables, high quality Cardas CABD banana were used to parallel the two cables. 10 AWG shorting wire plus spades were used for the “short” open-short impedance tests.

The test configuration allows a reliable and consistent termination. In-use would use the Cardas CABD banana and terminate the second cable spades onto the CABD as shown below. This removes a banana needed for the test set-up when we used locking banana on one cable end, versus the CABD, and spades on the opposite end.

The picture below shows the Cardas CABD banana terminated directly into the amp 5-way binding post with the second parallel cable spade terminated directly into the CABD body.


The following tables show the measured data for four sets of different cables both single and parallel. Each leg is measured for consistency, and then we parallel and re-measure to evaluate the theory to application. Four cables have been used to give a better idea how this potentially translates to other cable.



DCR METER Valhalla 4176

1313A – 10 feet. 10 AWG dual wire ZIP CORD style.

1310A – 10 feet. 14 AWG x 4 legs STAR QUAD style.

SERIES I ICONOCLAST – 10 feet. 24 x 24 AWG  weave style.

SERIES II ICONOCLAST – 10 feet. 48 x 28 AWG weave style.

The data follows the theory pretty closely. We see that R and L roughly HALVE and the C roughly DOUBLES in parallel. What can we do with this information? It is often difficult to drive inductance low in a single cable. If we chose a cable with the capacitance in check (100 pF/ft or less) we can consider the possibility to DOUBLE bi-cable arrangements and enhance the electrical much more economically. This is a nice way to boost reasonably priced cable’s electrical.

Two 1310A when attached to a single set of binding posts measure to very good electrical values in the table above. What if we want to bi-cable and bi-wire a double set to EACH set of speaker binding posts? The amplifier will see the total capacitive load for all four cables. This is why LENGTH and capacitance, both, need to be considered. Capacitance is per foot. For high quality cable parallel wire is a VERY good way to improve R and L electrical for speaker cable. It is extremely difficult to make a single aggregate cable that measures this well.

1310A when wired star quad, is two legs in parallel, and then for bi-wire (separating the woofer and mid/tweeter sections) two of those are in parallel to each amplifier’s binding post like the initial pictured example using ICONOCLAST, that’s four cables in parallel off each binding post to the speaker.

All the current Blue Jeans cable has low capacitance and tested in 10 foot real world assembly lengths and with connectivity. Why worry about capacitance when we all know that the first order filter roll-off is way above audible? Amplifiers that are too capacitive-loaded can oscillate. Modern amplifiers are better stabilized than wide-band amplifier of the past but still, adding a cable load PROBLEM that we need to solve with the amplifier’s design or a Zobel network isn’t the best engineering. This is why I have limited the capacitance of the cable to proper values that even pretty picky amplifiers won’t have issue with in 10 feet parallel, and even double parallel, lengths. Do check with your amplifier’s capability to drive the TOTAL capacitance.

Looking at the data charts above for each example the INDUCTANCE and RESISTANCE do drop to roughly HALF of what they were before. If capacitance isn’t too high, we can really give the cable assembly an electrical improvement with parallel cable. What does it do that we can see in the data? We have graphs for that.

All the data is real, and tested, with actual assemblies. Raw data is in the appendix support section. The first graph is the measured IMPEDANCE for ONE cable. Next graph is the IMPEDANCE after we wire two cables in parallel.  The data shows that parallel cables reduce the impedance substantially.

The graphs above are the OPEN-SHORT impedance of the ten foot samples. Audio isn’t RF, and we can’t test short lengths accurately other than open-short for accurate results as the wavelengths are too LONG to fit enough into the cable to be a true transmission-line (typically 10 wavelengths for a stable RF situation). The LENGTH has to be typical of the use as well, so ten foot assemblies have been used to keep the data comparable. Different lengths WILL change the data but not the pecking order of what’s low or higher impedance as the R, L and C are per unit length.

The IMPEDANCE is MUCH lower when we put two cables in parallel to the speaker terminals but passive cable cannot be eight ohm through low frequency audio as Vp, velocity of propagation, drops as frequency drops and RAISES the impedance. We can trick that rising impedance problem by doubling up cables and to better match the speaker load.

The frequency range to which the most power transfer function is being applied is the WORST impedance match to the speaker. Many discussions about cable center on this issue, and how it helps or hurts the sound quality.  Mitigating it (as best we can) seems to be the most appropriate answer, and we can budget the improvement for our needs.

1310A using star quad wiring and ICONOCLAST lower the impedance even when using single wire over 1313A “zip cord”. Dual wire lowers impedance and resistance even more. Zip cord type 1313A isn’t electrically ideal enough to mitigate the impedance issue when comparatively measured. Basic 1313A zip cord design cannot reach the better electrical. 1310A, a reasonably priced cable, eclipses 1313A when it is wired star quad.

The following two chart traces are just the loop path SHORTED component of the IMPEDANCE data. The 20 Hz and DC loop DCR figures should nearly match if the two test instruments are calibrated properly. I use the Valhalla for DCR and the HP unit for swept points frequency data. We do indeed see proper DCR and 20 Hz loop resistive values at the low DC/20 Hz anchor point.

We’d like to see “zero” resistance across frequency and what we see is a composite effect across frequency that increases the resistance even in the audio pass band. Skin depth, proximity effect and attenuation effects to name a few are the culprits. Please look at the UNITS. It is in mill-ohms. That a SMALL measure but it is measurable through the analog audible range and below even 10 KHz. 1310A and ICONOCLAST are far flatter through audio. Not all cable can measure really well here.

Series II ICONOCLAST does raise the capacitance on purpose to substantially reduce the Vp through audio, and this also keeps the impedance lower. Audio is a trade-off and you can’t have both at the same time- one affects the other. Series II flattens the VP so it is more equal through audio but the equation required a higher DCR (28 AWG insulated wire) and higher capacitance to do that. The provided low frequency Vp equation tells you what needs to happen, not how to make it happen.

Simpler 1313A and 1310A cable designs cannot do this optimization with fewer fatter conductors. The BIG wire’s DCR is too low.  Series II does have the best Rs, lowest impedance and lowest Vp linearity but it is designed to be paralleled to REALLY shoot ahead in the bass region to where it is then the best of the best everywhere.

Comparing the following two charts below we can see that the resistive component of open-short impedance is cut theoretically in HALF when we parallel the cables. The data also shows that that’s what we really see in the actual application. This is the BULK frequency resistance, not each individual insulated wire loop frequency resistance. All traces converge at the “DC” loop value at the left side of the charts.

Three designs have a lower impedance average and better resistive Rs uniformity than zip cord 1313A can offer, but the designs get more complex, especially ICONOCLAST that leverages multiple small wires for Vp properties that also optimally tune the capacitance limit at the low end to keep impedance low. ICONOCLAST uses 24 and 48 wires in parallel in each polarity. Then we put those in parallel! Yes, it is a complex circuit best measured as those values are the facts in real world use.

See the ICONOCLAST Vp tech paper. It takes a lot of design work to eke out the Vp effects and not raise the L and C too much, all the while keeping bulk DCR, swept resistance, and impedance low.


Using this data, we see that paralleling a REASONABLE length cable can benefit electrical performance. All this complexity is still subject to audible evaluations, as we just covered graphs and numbers to support the cable arrangement. BAV and ICONOCLAST are all about the numbers supporting what we do and why. If the supporting numbers aren’t better, what is?

APPLICATION ONE – for a VERY economical solution just parallel your existing single post speaker cable. This makes even 1313A look much better. If we are still wanting to better match speaker impedance at the low end (below 200 Hz or so) we can consider a more complex cable like 1310A or ICONOCLAST in parallel. Series II ICONOCLAST was designed to be used in parallel. Why? Because it eliminates the higher 24×28 AWG DCR. A more complex single cable to mitigate that higher CMA value would cost far more than using two Series II ICONOCLAST in parallel. The 65 pF/foot capacitance was purposely used to hold the impedance low as well as the total capacitive load.

The series I or II in parallel double the CMA. CMA, Circular Mil Area, is just the wire diameter squared and added up. When we add the wire path length and connectors compared to the bulk CMA DCR, we measure 1.185 ohm/1000′ and 1.49 ohm/1000′ for the series 1 and II respectively. See the data charts above and raw data.

1313A will be closest to the calculated CMA DCR. There is one single wire path length, so we measured 1.12 ohm/1000 feet.

APPLICATION TWO – Use a star quad like 1310A. This is really a MULTIPLE parallel “quad” situation. We are already technically two legs parallel with ONE 1310A star quad. This is how we drive down the INDUCTANCE compared to 1313A or zip cord. Put TWO 1310A in parallel and we have another doubling of the paralleling property. This is how the 1310A gets to where ICONOCLAST is on low frequency impedance numbers. For a really nice cost center, use 1310A paralleled to the woofer and mid / tweeter in what I call parallel and bi-wire assemblies.

APPLICATION THREE – This is the most elaborate and expensive method with any cable. Optimally we use ICONOCLAST series I or II. The series I can be used parallel in the bass region at a lower cost, and the series II parallel in the upper mid/tweeter. This is what you see in the earlier picture. Or, you can use series I or II in both places. For the best performance keep the series II ICONOCLAST paralleled in the upper frequency range.

NOTE – DO NOT parallel different cable designs to the same driver(s) as the time-based properties need to be crossed over between cable groups and through the frequency range each cable is used across.

SUMMARY Proper design principles will transfer to other applications if it is a solid, repeatable, process. We do have unique DESIGNS but the underlying physics is impartial, it works everywhere you use it. This is why 1313A and 1310A act the same as ICONOCLAST when paralleled

If a cable gets too complex to improve, why not take advantage of measurable benefits of parallel cable? True, we need to add-up the capacitive effects but most cable with less than 100 pF/foot capacitance and shorter lengths will allow you to try this measurably improved solution. Not all cable will show the Rs improvements 1310A and ICONOCLAST do, so be aware that DESIGN influences that parameter. And, the use of multiple small wires to flatten Vp and tune the low frequency impedance, both, are not possible in simpler designs like 1313A and 1310A because the conductors are too low DCR. I covered the Vp issues in a separate paper with calculated and measured impacts of what R and C do with multiple small insulated wire and why it is a better, but VERY complex, solution.

1310A, with the proper testing and certification for our hobby, does great job of bringing even more value that everyone can afford. Blue Jeans is happy to push the lower priced products as close as we can to ICONOCLAST. The provided data demonstrates that 1310A wired in a star quad and parallel arrangement does exactly that, it moves above and beyond 1313A but, it is a more expensive design. ICONOCLAST trickles down as much as we can.


The following are the actual test reports on each cable for those that want to see everything. One thing to note, that Rs impedance at 20 Hz has to near match the DCR tested with a Valhalla unit. The HP is a SWEPT frequency point set of data, and I always check that the DCR and the 20 Hz HP unit are in close correlation to verify the accuracy of the tests.

Shields and Grounding

BACKGROUND: There is always discussion on how to ground a shield. The answer lies in what the worst-case noise situation is. It isn’t always the same answer. Do you leave one end open or ground both ends? If you ground only one ends, which end?

We can look at how the interference behaves to answer these questions and they follow understood electrical properties within swept frequency regions.

BODY: The first step is to understand what a shield is doing, and how. In the simplest terms, a shield creates two separate “electrical” environments, one on each side of the shield. One side is measured as a RATIO of the field’s intensity relative to the other side, in dB.

The shields we are working with are ONLY effective with an EM wave that is predominantly “E” field in nature versus “B” field, or magnetic. Magnetic fields are not shielded or blocked with conductive shields, but need a shield material that blocks magnetic flux lines or a low permeability material…think “material a magnet will stick to.” Those kinds of material allow magnetic flux lines an easier path than through air. We can capture and re-route the flux lines in low permeability materials. For this discussion, we will look at mostly electric field shields, stuff that conducts electricity.

What is an ideal electrical shield? It is a shield that 100% blocks electrical energy at the surface of the shield, and that has infinitely low resistance. Shields don’t have infinitely low resistance and they block electrical energy at differing impedance based on the skin depth of the shield at a specific frequency.

We can measure the effectiveness of a shield across frequency with a transfer impedance plot. This is measured in milli-ohm/meter. It describes the resistance we can expect a shield to have, and thus the ratio of the fields energy in the shield based on the CURRENT the resistance causes to flow at that frequency…and it is not linear.

The graph below shows the frequency-dependent nature of a shield. A perfect shield would have NO RESISTANCE and both ends would be identical and thus seem like a SINGLE point of reference to a flow of current. Since we have zero resistance across the shield, we can’t have current flow caused by the shields. We CAN have current flow between the two points connected at the ends of the shield. In a “perfect” world the GROUND at both ends is the same potential and thus, an ideal shield has ZERO current flow, and is a measure of the POTENTIAL on one side of the shield relative to the other, in dB.

Since we don’t have a perfect world, Transfer Impedance describes what to expect at frequencies based on the shield’s impedance, and how that shield resistance creates a CURRENT flow and thus a voltage (shield resistance times the shield’s impedance = a voltage). When we have different resistances at each end of a shield we have current flow.

Shield Type5 MHz10MHz50MHz100MHz500MHz
Bonded Foil +60% braid2015112050
Quad Shield 60% +40% Braid
Tri-Shield+80% Braid10.60.10.22
Bonded Foil +95% Braid10.50.080.091

The chart below is what JUST the shield impedance looks like for a set length of cable at lower frequencies. We see the same non-linear behavior of shield and frequencies.

A SEED (Shield Effectiveness Evaluation Device per IEC 61196-1) test shows the dB relationship to a Lower Shield Impedance. Series number 5 with a 95% coverage 45-degree braid and Duofoil tape is clearly superior.

Okay, we can see a shield is not perfect, and not linear. So how does this say what to do with each end of a shield? We have to weigh the CHOICE of HOW the shield WORKS to decide our fate.

–     If you have ideal grounds and meet IEEE bus bar grounding (see the 568C.2 or later grounding specifications) limits it means BOTH ends can be grounded and the shield current will inductively couple less interference than the shield ATTENUATES through its material composition.

–     If we have severe ground differential, we can induce a strong current in the shield that CAN, if the shield’s resistance is higher, induce noise into the core that is WORSE than if we disconnect one end. We convert our shield to an antenna, not a shield!

–     An antenna does NOT create two separate environments between them with the ratio of one measured to the other. One end of the antenna is infinite impedance (the open end) with the other end at ground. The antenna “wire” is as close to zero impedance as possible in order to NOT attenuate the antenna’s signal going to ground. The signal won’t go to the open end, but seeks the lowest potential in the circuit.

We trade the noise caused by a poor ground resistance potential between shield ends for the induced noise in an antenna’s wire parallel to the signal wires that induces a voltage based on the antenna resistance. In an antenna type ground, it is best to ground the SEND end, as the SIGNAL on the internal wires is as LARGE as possible relative to the antenna signal, improving the signal to noise between the two.

An often-ignored aspect of shields is HOW to ground one at lower frequency versus RF. There is a big difference and again, it is based on the shield characteristics at each frequency.

The charts below are derived from TWO slightly differing MODELS of RF shield inductive reactance resistance. I have this paper for those interested. But, the data is the same message in that as frequencies increase, the shield reactance goes UP. This necessitates a FULL 360 ground at the shield termination point in RF circuits. This is why good RF connectors are fully capturing the shield all the way around the cable. On your RF digital cables, use 360 degree grounds for the best true shielding.

SUMMARY – Most systems will have proper GROUND differentials between them and thus have near ZERO shield current. The shield relative to the signal wires will ATTENUATE outside interference. When you have poor grounds, it may be beneficial to unhook the “receiver” end of the shield and hope that the induced antenna current voltage is less severe than the induced voltage caused by differential shield ground potentials. This should be a SECOND choice, not the first. A properly working shield, by design, has a KNOWN shield dB rating that can be trusted in a proper electrical circuit.

An antenna ground’s induced voltage onto the cable is not fully described and is dependent on the GROUND proximity point and shield’s distance from the signal wires. In severe situations, it may be the best choice to mitigate noise to the lowest possible reference value as it is pretty hard to REMOVE a shield already on a cable. Some, such as coaxial cables, can’t be mitigated and need to be properly designed to EXCEED the ground differential by several orders of magnitude so as not to aggravate any ground differential.

ICONOCLAST will use double ground interconnect shields and proper DCR RCA grounds. Power cables should also use grounds at BOTH ends if you have a proper GROUND plane resistance such that ZERO current flows and thus you have ZERO induced voltage from differential current. An antenna type ground CREATES a differential in each end of the “antenna” by design (one end is ideally infinity the other is ideally zero) and is thus a second choice if you have known ground issues.

One last note, those heavy 10 AWG power or more cables, may provide benefit as they induce less ground differential resistance than smaller power cords as the ground wires are larger. The circuit may not need the power delivery of a larger cable, but the lower ground resistance values may be of benefit on longer runs in marginal power grid situations. This will improve a shield’s current to nearer zero across frequencies. The dB isolation numbers values are for a proper shielded system with IEEE and TIA compliant shield differentials.

The Transfer Impedance numbers are between two-reference point probes on a shield, and DO NOT need the ground potential differences for characterizations.

BAV Cable — Design Notes

(This article consists primarily of a technical piece by Galen Gareis, supplemented by some paragraphs by Kurt Denke, as noted herein by the initials “KD”)

BACKGROUND: The development of the ultra-low R, L and C ICONOCLAST™ home studio XLR cables exceeded design expectations and performance numbers reached. The patented use of true air tube dielectrics enabled electrical performance relative to size that is unattainable any other way.

The drawback to the pure performance ICONOCLAST™ design is that the ultimate electrical performance does require a mechanical limitation in the cable’s use that is wholly managed in the home audio space as performance is needed above all else, and in a more controlled and fixed installation.
The move to a STUDIO and professional market seems nearly impossible to reach the same size and electricals as the home market design. This paper covers the unique and high performance electrical XLR design that also meets, and exceeds electrical expectations for such a strong, flexible, crush and impact tolerant design.

(KD: While this paper does not address the BAV RCA cable directly, I will append a note at the end to tie those in.)
BODY: The most difficult task was the design move from foamed Teflon with solid Teflon belting in the original design to a MUCH more ruggedized version for the studio market. The careful design of the AIR TUBE dielectric used for home use cables improved the cable’s air tube physical performance (resisting deformation and physical changes under use and to advised bend limits). Changing the process variables (bonding the outer tube to the inner filler) and materials (use a modified POE, Poly Olefin Elastomer) allowed a remarkably tough, flexible and crush resistant core structure. The POE’s electrical characteristics were mitigated, as was the Teflon’s in the home ICONOCLAST XLR), by putting AIR between the conductor and the solid dielectric materials. Where the dielectric materials were required to touch the wire the lowest dielectric material was kept in place, Teflon)
The selection of materials was important on two fronts: modulus of elasticity and dielectric properties. If either was deficient, the design would not work.
ELASTIC MODULUS – The modulus of elasticity (also known as the elastic modulus, the tensile modulus, or Young’s modulus) is a number that measures an object or substance’s resistance to being deformed elastically (i.e., non-permanently) when a force is applied to it. The material has to stretch or compress but with a high force (lower modulus) applied. This is called elastic deformation.
The chart below shows the graphical properties of plastic deformation. We need to work in the LINEAR region so the part will recover back to its original shape. – definition.
Linear elastic deformation is governed by Hooke’s law, which states:
σ = xE
Where σ is the applied stress, E is a material constant called Young’s modulus or elastic modulus, and x is the resulting strain. (This relationship only applies in the elastic range and indicates that the slope of the stress vs. strain curve can be used to find Young’s E). Engineers often use this calculation in tensile tests. The elastic range ends when the material reaches its yield strength. At this point plastic deformation begins.

In order to resist plastic deformation under tensile and compressive load simultaneously, a material with a high elastic modulus would be best. The material selected was DOW ENGAGE 8450:

The new part had to meet the same SIZE requirement for XLR compatibility as the TEFLON part, so material properties are extremely important. The PHOTOs below show the TEFLON part on the left and the DOW8450 part on the right.
A very important design change other than the material was to BOND the outer POE material tube to the inner POE filler. This required a SOLID material to also enhance the YOUNG’s modulus of the overall part. Bonding the two parts, the filler and tube, is much more critical than it seems for the intended application of this part. In order to meet crush and impact resistance with the part, it had to be bonded to SIMULTANEOUSLY be crushed and stretched concurrently.
When a tube is crushed, the sides will EXPAND in dimension. The inner filler X-design provides a high degree of tensile resistance to this elongation change, improving the part’s deformation resistance. Once the force is removed, the part pulls back into its original shape. This is why a high 1060 PSI material is needed with a LARGE elongation percentage: 750%.

Left: Teflon, no interface bonds; Right: POE, with interface bonds

The pictures below illustrate the TUBE’S properties under compression. With no filler in the center, the tube will crush as easily as the material properties allow. To SIGNIFICANTY improve the crush values, an inner BONDED member is added across the lines of force.

The inner member resists the expansion of the part due to compressive loads. The above “POE PART” adds TWO cross members for the air channels and strands the cable with a helix lay so under compression there is always a cross member restricting the parts expansion under load.

In the above example, the cross member goes under TENSION when the tube is compressed. This HOLDS the tube’s deformation in check, and more importantly, it has a MEMORY to what it was prior to crush, and the high elastic modulus pulls the part back into the original shape under normal compressive loads. The cable “heals” itself back to its original form. The core is tightly stranded to make sure there are always a cross member perpendicular to the load vector direction and / or TWO members largely perpendicular to the load vector.

Part is tightly STRANDED to place X-web in ideal location for compression performance and recovery.

When a part is tightly bent, it is under COMPRESSION (inner surface) and TENSION (outer surface). This hybrid part is ideally made to resist this type of plastic deformation, and recovery.

This hybrid part is ideally made to resist this type of plastic deformation, and recovery.

Since we had to use a SOLID material for ultimate Tensile / Elongation properties of the part the dielectric properties are going to be impacted to some degree. Is the erosion in electrical properties going to allow the overall high electrical standards that are also required of the finished part?
Material samples were taken, along with the TEFLON reference and measured for dielectric properties. The tables below show the excellent measured electricals relative to TEFLON. We should see minimal electrical impacts in the Audio frequency range with the exception of CAPACITANCE which is directly tied to the dielectric material. But, the use of AIR TUBES mitigates the changes of capacitance.

POE is ~2.4 versus Teflon ~2.1, a 14.3% change.
POE and Teflon share the same dissipation factor.

The capacitive effect is logarithmic so the farther the material is from the reference metallic member the lower the capacitive effects. Twice the distance is one-fourth the effect. We have AIR inside the part, which is the best dielectric there is. Where material does contact the wire, we use a TEFLON thread to mitigate the capacitive effect of the wire, to thread, to outer POE material tube.

Where the magnetic fields are strongest, near the wire surface, we need AIR and/or minimal dielectric influences (use TEFLON thread). The outer influences decrease significantly with distance. If a surface is infinitely away from the conductors, the electromagnetic effects are zero. The use of AIR as a dielectric (constant = 1) allows “infinity” to be reached much sooner than if higher dielectric constant materials are used between ground plane (conductor to inner braid surface). The task is how to do this and allow extreme durability and flex, while meeting superb measured electricals. The careful positioning of material is necessary.

Air-spaced conductors, with FEP Teflon thread spacers

The overall GROUP delay is the superposition of ALL the material between the conductor and outer braid. The use of AIR keeps the VP and capacitance changes in control per measurements below on a braided core.

Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture:
Cap @ 1 kHz Spec est: 8.0 +/- 2.0 pF/ft nom. 60485Y – 8.95 pF/ft

Inductance @ 1 kHz per ELP 424, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture:

Inductance Spec: N/A (uH/ft) 60485Y – 0.15 uH/ft

Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed.

VOP Spec est: 87 +/- 1 % 60485Y – 84.7%


Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Cap @ 1 kHz Spec: 10.5 pF/ft max PDC2882 – 10.1026 pF/ft
Inductance @ 1 kHz per ELP 424, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Inductance Spec: N/A (uH/ft) PDC2882 – 0.1542 uH/ft

Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed.

VOP Spec: N/A (%) PDC2882 – 83.4%

The above tested numbers are excellent compared to the CONTROL all FEP design. The estimate capacitance is negligible with nearly the same group VP:

Super Flex Mic CableIconoclastLLDPE Design
8.95-10.5 pF/ft (pin 2-3)5.5 pF/ft9.44 pF/ft
0.15 μH/ft0.153 μH/ft
83.4-85.4% nom VP87%84.2%
Dia. 0.350″0.325″0.350″

A cable can be crush resistant and impact resistance but STILL be too inflexible for the intended use. This is called the BENDING MOMENT. Or, the amount of force to bend the cable ninety degrees around a specified reference mandrel when comparing cable types. This cable is EXTREMELY flexible and bends ninety degrees under its own weight (illustration).

SUMMARY: The PDC2882 initial design trial has been extraordinarily successful at meeting electrical and physical design parameters. The design actually well exceeds expectations using the DOW ENGAGE 8450 material properties.

The PDC2841 LLDPE design met similar electricals, but fell far short on physicals.

We can run a full extrude jacket with paper tape separator to improve cosmetics (perfectly round) and enhance crush even more.

CONCLUSION: The DOW 8450 material design is the best electrical and physical XLR cable design for heavy duty studio application with extremely high audio standard properties for sound quality. The DOW 8450 inner core member should provide industry leading sound quality with the best durability possible. The cable’s self-healing tensile modulus properties insure a higher degree of user abuse than any air core cable made today.

  • Best dielectric near the wire is NO dielectric.
  • We use AIR tubes, and tubes are usually STIFF by design.
    o 25 AWG signal wire insure high flexibility with a tight stranding lay.
  • A special physical and electrical material was sought after and found for this design.
  • Our BONDED perimeter air TUBE core solves the CRUSH and recovery issue.
  • Air tubes are now an ADVANTAGE in a flex application.
  • The target electricals were to be as near the all Teflon ICONOCLAST version as possible.
  • We met that goal and with FAR, FAR superior physicals;
    o Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
    o Cap @ 1 kHz Spec: 10.5 pF/ft max PDC2882 – 10.1026 pF/ft
    o Inductance @ 1 kHz per ELP 42C, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture
    o Inductance Spec: NA (uH/ft) PDC2882 – 0.1542 uH/ft
    o Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed.
    o VOP Spec: NA (%) PDC2882 – 83.4%
  • Fits Standard XLR with a 0.350” outer dimension.
  • Cable exhibits standard XLR benefits of reduced external noise with passive CMRR design.
  • Air dielectric insures the best possible conductor interface.
  • Smaller signal wire insure better signal coherence (more even current distribution with respect to frequency)
  • Star quad design double-up the CMA are to act like a larger wire for longer distances.
  • CMRR keeps ingress noise low, even magnetic origin, with a tight strand lay (even exposure to external noise).
  • 90% BC braid lowers EMI / RFI noise BEFORE it encounters CMRR on the four signal wires.
    o Wire should be purely resistive for time coherence at audio but it isn’t;
    – Low capacitance keeps reactive first order filter roll-off values well away from the audio band.
    – Low inductance improves reactive current related variables to a minimal reactance.


Supplemental Note:

The paper as written, above, by Galen Gareis does not directly address the RCA cable design, but the rationale is very much the same. As Galen sets out in his Iconoclast RCA and XLR design briefs, the electricals for the RCA really ought to track those for the XLR, and in the case of the Iconoclast product they do — the RCA design is very much like a quarter-section of the XLR, eliminating two of the four quarters because the cable is unbalanced rather than balanced, and the last because the purpose of paired conductors on each polarity in a star quad is lost when common mode noise rejection is lost, as in the transition from balanced to unbalanced.
This same design principle has been followed in the BAV product. The BAV RCA has a single air tube and filament, like one chamber of the BAV XLR. The resulting characteristics track those of the XLR cable as closely as possible, avoiding any characteristic difference between the sonic qualities of the two cables. )

Speaker Cable Design Brief

Speaker cables are a very different animal than high input impedance interconnecting cables. A speaker cable connects to an extremely inconsistent 2-32 ohm (or even lower and higher!) reactive load created by the speaker. RCA and XLR interconnect cables see a much more consistent and resistive high impedance load making their electrical measurements far easier to predict. The speaker cable also suffers from the audio band’s velocity of propagation non-linearities seen in the interconnect, but also has to figure out a way to be LOWER in impedance to better match the speaker load, while the velocity of propagation is going DOWN, and this naturally increases the cable’s impedance. How is all this managed as best we can?  This paper is a walk through on how ICONOCLAST™ speaker cable addresses some of these issues.



  1. Conductors.
    1. Copper Size.
  2. Dielectric material(s).
  3. Dielectric geometry.
  4. Shield material and design considerations.
  5. Jacket design and material considerations.
  1. Conductors

For speaker cables, the first issue that has to be decided is how much CMA (Circular Wire Area) you need based on the application. This isn’t always an exact science as the cable length and speaker type will change your calculated answer. The speaker cable becomes part of the cross-over network in the speaker. The amplifier sees BOTH components as one load. 

Since the cable is seen as part of the speaker, it is easy to understand that the “reactive” relationship is between the speaker plus speaker cable and the amplifier.  Speakers vary by design so the overall speaker component back EMF portion of this load into amplifiers varies. Amplifiers of differing design react to the back EMF and the overall performance can be hard to predict. The goal is to “remove” the cable as best we can between the amplifier and speaker. Cables should not be tone controls, but that’s the goal of EVERY component!

The analysis below looks at the calculations that have been made to settle on the total CMA area for benign reactions to the frequency response of a typical set of loudspeaker loads. And yes, these are NOT real time resistive loads but as always, an approximation.

The general rule of thumb is that you want the total speaker cable resistance to be less than 5% of the speaker impedance PLUS the cable resistance value to avoid speaker frequency response interactions;


Vout = Vin x R2 / (R2 + R1)

ICONOCLAST™ total CMA size mitigates appreciable calculated frequency response changes, and stopped at 9600 CMA (10 AWG). 

For most practical applications of 0 to 35 feet, 9600 CMA per polarity should work well to be resistively invisible to the speaker, or amplifier. We want the load to be the speaker, not the cable.

HOW we get to the approximate 9600 CMA per polarity is the hard question. For those that want the easy way out we have one, 1313A. If we want to see if we can DESIGN a better MEASURING cable let’s see what can be done with Belden technology.

In order to figure out what best to do, I looked at things that indicate what NOT to do. We all know by now, that multiple smaller wires (to a point!) are better than one fat 9600 CMA solid or stranded wire.  The operative here is, to my ear, TIME based issues at audio. You want the signal to be most uniform through the wire for improved current coherence (more identical frequency arrival times). To make that happen, we decrease the wire size so that the skin depth penetration goes deeper into the wire, evening out the differences in current magnitude with respect to frequency.  This technique better aligns the signal speeds through the wire. I said “better” as there is no perfect way to do this. But we can certainly be better. The depth is calculated based on frequency and material. The wire size does not change the penetration, it DOES change the minimum current found in the center of the wire. The smaller the wire, the closer the center current magnitude matches the surface current as signal frequencies go up.

Studies were made on various geometries that would hint at what type of conductor to use, and how many. What various design limitations be “inside” the ~9600 CMA resistive box we want to be within?

Probably the easiest approximation for a cable with multi-sized wires is a flat design.  Yep, line those wires up and stop when you reach the proper AWG size.  The parallel wire tested issues lead (pun there?) me away from this simple design. Why? I looked at our TEFLON® ribbon cable for that answer.

A really nice “flat” TEFLON® Ribbon Cable

Above is a TEFLON® ribbon cable I used to test polarity symmetry, and capacitive symmetry WITHIN each polarity.  The two tables below graph the capacitance from the outer edge wire to the opposite polarity, all opposite polarity wires grounded together.

1 kHz10 kHz
STD DEV=0.8620.761
@1 kHz

The data says that the CONSISTENCY of FLAT cable is not perfect. The closer each wire gets to the opposite polarity, the higher the capacitance. The GROUND reference is more robust the closer we get, and the less distance between two wires, all else the same, the higher the capacitance. We have EACH and every wire, for all intents and purposes, acting like a different wire. ANY cable with more than ONE wire per polarity will have this issue to contend with. How can we do better on capacitance control in each polarity?

For the answer to that we need to turn to inductance. When you separate the two polarities in a flat design, inductance is seemingly well controlled. Each parallel wire has current going in the exact same direction in each polarity half so the magnetic fields CANCEL one another. The closer to the inside polarity separation zone you go, the more the opposite polarity’s different current direction upsets the SYMMETRY of the inductive cancellation process. There is non-linearity through the “flat” polarity, too, but it is worse near the edges of each polarity where the “design” changes.

Two wires with the SAME current direction next to each other cancel some of the fields’ gauss density between them, and two wires next to each other with opposite polarities reinforce the magnetic field lines.

Below are two close proximity wires. Notice that the current direction “adds” between the wires with the magnetic field flux lines in the same “reinforcing” direction. If we FLIP the current direction of one of the wires, the currents cancel but now we have two of the same polarity to get the cancellation effect. This is the problem with ZIP cord. We can get low capacitance, but it is not practical to get the lowest inductance.

To prove a point, a single bonded pair used in ICONOCLAST measured by itself is 12.5 pF/foot and 0.196 uH/foot inductance, about what 1313A reference zip cord is (chart below). This isn’t the best reactive variable balance of L and C for a premium current delivery cable.

In the tested flat design there are inconsistent ground plane issues that have to be resolved, AND there are inconsistent electromagnetic field cancellation properties, too, through the “flat”. The problems are locked-in by the geometry of this cable specimen, same as the issues with zip-cord.

What is GOOD about a flat cable that and can we use those positive attributes and mitigate the bad aspects? The answer to that question lies in a BONDED pair used at RF frequencies. To get to the answer for speaker cable, we need to re-invent what a BONDED pair does at audio. Re-designing a bonded pair for audio leads to what size and count wires we can manage in forward processes. We STILL don’t have the conductor size or quantity question answered after all this.

What is a bonded pair? A BONDED pair is two co-joined wires. A super geometrically consistent zip cord design with superior adjacent wire BOND technology. The precision C-C of each wire controls impedance at RF to incredibly small variation.


A zip cord removes a lot of symmetry complexity for poor magnetic field cancellation properties. Adding wires to the zip cord to make it a FLAT cable just adds to the capacitive and inductive “cable in a cable” issue as every wire becomes its own drummer. Coherence is improved with more small wires that add to the same CMA, but we don’t really have “one” like polarity for each signal anymore.

Tests show the inconsistent capacitance in a FLAT arrangement. Tests can also show the INDUCTANCE issues with zip- cords. A single bonded pair is 0.196 uH/foot inductance. This value is far too high for the state of the art R, L and C cable that is the intent of the project.

How is using another bonded pair zip cord component going to fix this mess? The answer is in the XLR cable. We need to build STAR QUAD arrangements of BONDED pairs! Visualize the currents using the right hand rule;

Like the XLR, two BONDED pairs in a QUAD arrangement show ideal field cancellation with LIKE polarity current all in the same within the same polarity. This field cancellation property of star quads tells us fundamentally we need two polarities using many wires in a star quad arrangement. There isn’t an answer as to how, yet, just that a true star quad is a key element we need to keep.

The solution was a compromise, as is usually the case in audio cables. The design devised a way to create star quads THROUGHOUT a process that varied between near perfect, and slightly imperfect. It was done with 100% consistency within each polarity so every wire measured the same inductance and capacitance to the opposite polarity, and made significantly lowered inductance with only a moderate rise in capacitance. The capacitance was increased on purpose, I might add! More on why I did that later.


The above illustration shows the variation in the STAR QUADS between like bonded pairs in a polarity. The question is does it work; capacitance measured 45 pF/foot between polarity wires and inductance measured 0.08 uH/foot. Capacitance variation, and the electromagnetically tied inductance variation, is superb.


The difference in reactive stability between each wire in a single polarity, and BETWEEN each polarity can measures significantly better in ICONOCLAST.

1 kHz10 kHz
STD DEV0.1660.202

Tolerance is +/- 0.5 pF @ 1 KHz or more than 5 times tighter variation than the 8R28064 flat cable.

What was done was to BRAID, on a GHz capable braider, the needed wires to arrive at the 9600 CMA DCR requirement. The braider needed a symmetrical arrangement so an even number of bobbins was chosen, 12.  This is 24 wires per polarity. 9600 CMA / 24 = 400CMA per wire, or a 0.020” 24 AWG wire.

The braid DESIGN is not forthcoming, so the balance of electricals has to be understood.  Several, several design iterations were trialed before I froze the design around the proper braid relationship to arrive at a suitably balance reactive cable measurement.


People will “guess” that ICONOCLAST is a BONDED pair ETHERNET cable, and it is not. The REASONS and the DESIGN are not the same at all. All that is the same is the coincidence of a 24 AWG solid copper wire common to Ethernet.

Each polarity is BRAIDED and FLATTENED into a, you guesses it, FLAT shape! We essentially “fold” the flat cable over on itself into ONE polarity. Then, opposite polarities are tightly bound to keep LOOP area to a minimum, critical to inductance as the formula is GEOMETRY controlled, not the dielectric.


Measured Rs (skin effect / proximity effects)

            The nature of the magnetic fields can be indirectly MEASURED with an Rs measurement. The flatter the Rs, the better the skin depth / proximity effect are managed. Proximity effect is the currents in each polarity being “pulled” to the inside edge of each conductor, and away from the outside edge. This impacts conductor efficiency.


An awful lot of testing was done to identify the weaknesses of various designs. We wanted to avoid;

  • Inconsistent capacitance in each wire.
  • Inconsistent inductance in each wire.
  • Inconsistent ground plane interaction between wires and between polarities.
  • Inconsistent wire DCR between all wires.
  • Poor polarity DCR values (too high or low total CMA).
  • Inconsistent dielectric performance between each wire.
  • Poor frequency coherence in each wire.

After all the testing, a 20-mil wire diameter in a 24 wire (12 bonded pairs) woven polarity was created to match the design to the electromagnetic requirements.  The final design that drove the final wire size is 100% symmetrical in every measure on every wire.

Woven single polarities achieve class leading performance in polarity-to-polarity and wire-to-wire consistency while also providing exceptionally low reactive variables. The superposition of the magnetic fields drive inductance down from 0.196 uH/foot to 0.08 uH/foot, a 59% reduction in inductance, while holding capacitance to just 45 pF/foot. L and C can be CHANGED based on the woven DESIGN, but was optimized for speaker cable applications.

  • Dielectric material(s).

TEFLON® was chosen as it is again, the best solid dielectric there is. I needed a thin wall to bring the wires close together for inductance reduction but capacitance is an issue with 24 closely spaced wires. A capacitor is two parallel conductive plates with an insulator between them. To lower capacitance, I wanted a low dielectric constant plastic, Teflon®. To achieve the required low capacitance, more needs to be done to “thicken” the insulation without increasing loop area effects.

This seems impossible to do, but it isn’t with the woven design described above. The final insulation wall was driven by BALANCING capacitive gains with inductive reduction.  Dielectric geometry allowed this balance to be accomplished.

  • Dielectric geometry.

The requirement to meet capacitance ALSO drove the design to a weave pattern. Each polarity is SEPARATE from one another. There is NO interweaving of same polarity wires.

Some will ask about wires with several AWG sizes. Current will flow along the path of least resistance. This does not mean current won’t flow in specific wires, just that the majority of the current magnitude is shifted to the easier path. EVERY wire will have current at ALL frequencies. The magnitude will change and follow ohm’s law. Many differing wires sizes and electrical lengths can impact the signal arrival times across the audio band based on physical conductor lengths in composite wire size designs.

If we take two wires with the same exact skin depth (same frequency point being considered) but one wire has twice the surface area, more current will flow into the larger surface area wire. It offers less resistance. But, the lower resistance wire is a larger wire and isn’t what we would like if the current across the wire is to be more uniform. Bigger wires are better at lowering resistance at a given frequency because they have the most surface area. We use this at RF with a “skin” of copper to carry the lowest, yet still high, frequencies efficiently. The wire’s core under the copper is a material that is “filler” and has no current flow: steel, aluminum, etc.

At lower frequencies the current is diffusion coupled evenly through the ENTIRE wire. So if you send JUST low frequencies, use low a DCR wire as you can get.

Those are the extremes. Audio is weird in that we need to improve current coherence through the wire while it is trying to MOVE to the outside surface. We don’t care about attenuation as much at audio since it is negligible. We make the conscious decision to go for forced current coherence with more SMALL wires. This technically violates the practice of more “surface” area for lower attenuation at high frequencies for current coherence. Big wire is more surface area for attenuation while small wire is better current coherence but higher attenuation.  If you use one wire (interconnect) the current delivery has to be considered to the load. RCA and XLR cables have near zero current flow into the high impedance load so we can go for signal current coherence and suffer little attenuation. Speaker cables can’t use too few wires as there are 20-30 amps coursing through a speaker cable.

Audio is trying to TIME align the low and high frequencies, so the best, and most consistent, way to do this is to use more small wires that add-up to the low frequency DCR needs, and are small enough to FORCE the wire to see more and more cross sectional current usage at higher frequencies.  This means several small insulated wire that all need to be the same “single” wire.

The unique woven design does a LOT to reduce inductance and associated capacitance. How is 59% reduced inductance over a single bonded pair achieved?

    • Star quad wire arrangement.
      • Allows ideal geometry for low field strength.
    • Boned pair like polarity wires.
      • Allows star quads to be formed throughout the weave.
    • Separate polarity halve fields are NOT parallel, reduce field reinforcements.
      • Fields between polarities have some cancellation (wires that cross at ninety degrees cancel) since the cross at ANGLES, and not ever parallel.
  • Controlled Proximity effects / Skin effects
  • Measured Rs flat to 20 KHz.
    • Low dielectric constant plastic.
      • Thinnest possible C-C with the lowest cap.
    • Woven pattern averages out the wire-to-wire distances significantly.
      • Woven pattern separates the wires and “tricks” the bulk capacitive value to be far lower.

The last point on the capacitive reduction is also what we like in a FLAT design, but it is inconsistent. Average distance between any two wires in a braided polarity and thus between polarities is far more consistent.  The weave moves all the wires evenly, and consistently, to a closest proximity position and a max proximity position throughout the weave.  Capacitance and inductance DO vary, but they are exactly the “same” wire and at the same time as every other through the weave. The fattened weave holds overall capacitance to an unexpectedly low value of 45 pF/foot in a cable with such high conductor count. 

Low inductance leverages the same current direction in the bonded pair’s combined with the star quad wire geometry periodicity (end view photo above). And finally, the TIGHT textile weave between polarity halves force a low loop area and with wires never being parallel, further reducing inductance.

The overall reactance of the cable is shown in the graph below.

The chart illustrates a significant drop (yellow trace) in cable impedance compared to 1313A (blue trace).  We know all we need to know to figure out why this happened. The velocity, although variable, is nearly the same at each SPECIFIC swept frequency point. We need to look at frequency by frequency calculations. The capacitance is linear across the entire audio band so that’s a set value.

We have a set value of capacitance, and a nearly set value of velocity (there will be slight variation) at a given frequency. What is CHANGING is fundamentally the capacitance between cable designs for “impedance” characterization.

The impedance equation is influenced by the change in capacitance and thus lower measured impedance as the capacitance shows up in the denominator of the impedance equation.  Increasing capacitance from ~16 pF/foot to ~45 pF/foot decreases ICONOCLAST cable impedance. Speaker cables require low inductance and to get there without shooting capacitance through the roof. DESIGN is the overriding requirement, and materials alongside unprovable theory, are second.

Now we know why ICONOCLAST has the capacitance it does, as I can balance the inductance to industry leading values AND keep cap low, yet not so low as to increase impedance too high relative to the input requirement (impossibly low speaker impedance 8-ohms ideal). Cables go UP in impedance as you drop in frequency, the opposite of what we want. Listening test have to decide if the superb inductance or impedance matching with much higher cable capacitance is ideal.  Quick calculations will show capacitance problems with 8 ohm cables at audio once an amplifier is attached.

Don’t ignore the reactive time constants of L and C. We want an 8-ohm cable with NO L and C and zero resistance and you can’t do that. Getting cable “impedance” reasonably low is more reliably safe for amplifiers and TIME based distortions (lower L and C).

  • Shield material and design considerations.

I kept this topic here on purpose.  Some may already know that low impedance cables signal levels negate the need for a shield. And that’s a good thing because a shield over a speaker cable is darn near ALWAYS a bad thing for two reasons;

  • A shield will always increase capacitance of the cable. The question is how much.
  • To mitigate the capacitance increase, the shield must be moved significantly AWAY from the core polarities, increasing the size of the cable.

Shields are ONLY beneficial if the environment demands them. Shields inhibit the performance of cable in most cases. Coaxial cables being an exception as the shield defines the cable’s natural IMPEDANCE.  The ground plane proximity and uniformity are vitally important with short wavelength RF cables. Coaxial cables do just that. Audio is not RF, and these shields are more FUD devices than actual benefits, especially in speaker cables that have signals orders of magnitude over the background noise. Incidentally, the woven pattern in ICONOCLAST has a built-in immunity to RF not that that RF immunity is evident in the use of the cable.

View a SHIELD as a rain coat; great if you have water flying around but a major hindrance if you don’t. Audio seldom needs shielding on low impedance cables and here is why;

Magnetic fields decay rapidly with distance; ratio of 1/x^3. The best defense is to MOVE the low frequency electromagnetic cables away from one another. The foil and even braid shields are higher frequency shields that are ineffective at much below 1 MHz.  Magnetic fields lines need low permeability shield material (something a magnet will stick to) to route flux lines away from sensitive devices. A faraday cage is an example you can put something into to do this. Low permeability metallic shields are a pain to use (stiff and heavy). DISTANCE is the best remedy.

For EMI and RFI, the foil and braid shields used on Interconnect cable will be fine for RFI ELECTRIC field issues, but NOT 20Hz-20KHz magnetic fields.  Interconnect cables MAY have wide band input op-amps that can be needlessly hampered by RFI on the line. Speaker cable signal levels are many, many orders of magnitude above the RF and ICONOCLAST speaker cables aren’t a good RF conductor due to the weave pattern in the design.

  • Jacket design and material considerations.

All ICONOCLAST cables use FEP as the jacket to reduce UV sensitivity, plasticizer migration and chemical resistance.  The cables are designed to last decades.

SUMMARY – Little has been left to chance in the design of ICONOCLAST cable.  All the products are born from strict measurements and the management of known electrical parameters. Belden’s philosophy is to make as low and R, L and C cables as technically capable. The improvement to some may be unimportant. To others, and using different systems, they can be significant. The closer we manage the knowns, the better the tertiary elements will move along with those improvements. All cables “react” differently. ICONOCLAST is designed to offer the most benign interaction possible between your amplifier and speaker by leveraging high speed digital design principles to the much more complex audio band.


I have four 1 meter XLR sets going from the bottom component, up to the pre amplifier component. The natural LOOP needs to be allowed to form, unobstructed from the rear wall. This takes about 12″-13″ of clearance from the back of the unit to the wall. The rack may be closer to the wall than 13″ based on the depth of the rack. If the rack shelf is 24″, and the device is 19″, the rack will be only 13″minus 5″ or 8″from the rear wall, for instance.

Space the racks and devices APART helps, too, so a 180 degree loop back for one device on top of another is too severe. A bend radius of 8″ or so is needed (see the picture). If devices are such that the XLR are well offset, a piggy-back stack can work. The distance BETWEEN the XLR jacks is what is important, up and down or left and right or a combination of the two works.

The XLR will twist no more than 180 degrees to plug in for ANY wire path. With a loop-back, a change in the vertical alignment, to the left or right, will force a TWIST in the cable. Same with a straight through if the equipment is shifted left or right. Again, this is simply how XLR works.

Plug the MALE XLR in first. The female has a thumb tab that lets you quickly see the direction to go that puts the least torque into the cable. Simply rotate the female XLR to snap it in. All polarized plugs work this way. The BAV is designed to simply have a very low bending moment design that absorbs the exact same twist, but with less force…thus it can take severe routing.

ICONOCLAST used with adequate rear clearance can use 1 meter cables, but the FULL natural loop needs to form to allow the torque to be absorbed along the full length of the cable. This is worst case as the jacks are pretty much one straight above the others.

ICONOCLAST are true AIR tube with the best dielectrics available. These cables will require proper routing and rack positions. For those that can not conform to the reality of the design, I have engineered the BAV to be the best flexing (first requirement) and best performing for the flexibility. These will accept harsh routing.

Galen Gareis

iconoclast design engineer

Iconoclast Gen2 interconnect update

NOTE: This paper was originally written prior to the introduction of Iconoclast Gen2 interconnects, so while some references are to the future, that future is now….

BACKGROUND:  To possibly improve the performance of the XLR, to maybe achieve even lower L and C,  we would need to revise the current design…and it will jump up the electromagnetic complexity. The balanced of L and C would shift some but the coherence will improve substantially.

Changing the “conductor” to a four insulated wire structure will lower INDUCTANCE through signal phase cancellation. The star quad arrangement will retain CMRR for NOISE reduction. Four smaller wires will improve PHASE, and lower wire loop DCR to mitigate ground loops.

                                                                                                PROTOTYPE IMPROVED DESIGN:

Capacitance is the DISTANCE between the plates (wires) and dielectric material(s).

Inductance is two-fold;

  • The electromagnetic field cancellation.
  • The loop area between the wires changes inductance.
  • For inductance dielectric doesn’t really matter, inductance is DISTANCE.

We will have the same nearly loop area in the design (C-C distance is the same) but each conductor in the new design will further remove signal electromagnetic fields based on the cancellation geometry. Inductance should drop compared to the single wire conductor system.

The capacitance requires the same meticulous attention paid to as the group dielectric. Since we are keeping the cable the same size so the capacitance HAS TO go up as we have more wires parallel to a dielectric (the center X-filler, beading and outer tube) , and closer to the dielectrics. The conductor size and dielectric determine the final size. The added wires and X-filler close to the wires are the main contributors to the required capacitance increase. But, lower inductance improves PHASE shift, and your ear is most sensitive to.

The current coherence, the main objective of the design with minimal L and C changes, is based on the skin depth penetration changes going from 1 x 0.018” wire to 4 x 0.010” wire for each conductor.

4 wire “conductor”

Technically, four -wire per conductor will increase capacitance some as we have more wires parallel to a dielectric, but the  current coherence improves substantially, time aligning the low to high frequencies.

18214.4 μ inches = 18.2 mils @ one skin depth.

One skin depth is defined as when the surface current is 37% smaller going into the wire.  If we had a wire that was 18.2 mils in size, the CENTER of the wire would have only 37% of the current measured on its surface.

Skin depth equation (below) is a squared equation, so removing wire depth rapidly increases the inner current magnitude. Dropping from 20 mils to 10 mils is a 4X improvement in current coherence.

The very good, and easier to make, current design does NOT use electromagnetic signal field reduction technology I developed for the speaker cables in the series 1 signal leads. The current XLR design relies on reduced loop area and uses AIR to reduce the capacitance to a minimum for a given tighter spacing to achieve inductance.  The better the dielectric the CLOSER I can physically locate the signal wires for a given capacitance, thus lowering Inductance. The size of the wires determines the current coherence, and with more uniform effect of the dielectric around each wire with respect to frequency. The smaller the wire, the more uniform the velocity of propagation from low to high frequencies.

An XLR cable’s external noise utilizes CMRR based on all four noise signals being equal on each wire and which cancels those noise signals in a star quad design through electromagnetic field cancellation. If we look at the four wires, and using the right hand rule (current out of the page). All the external noise currents in the wire go CCW around each wire suspended in space. All the electromagnetic fields cancel adjacent to any wire and across from any other wire. All the fields superimposed onto one another forming a nearly ideal cancellation circuit. Nearly perfect because stray magnetic fields would extends OUTSIDE the four wires and reinforces the field. A first approximation says that this doesn’t happen. The stronger fields are closets to the wire and cancel most aggressively. Theoretical outer fields are weak, and don’t reinforce nearly as much as the inner fields cancel.



We DO NOT see this nearly “perfect” rejection of signal magnetic fields to reduce the inductance in the signal fields for series 1 RCA or XLR cable. We have a PLUS and MINUS balanced signal current direction whose fields are only partially cancelled. The partial field partial cancellation RAISES the inductance above “zero” theoretically as we have a stronger field, and separated by the distance needed to lower capacitance with any dielectric. The old design has ~36% higher inductance, and thus worse PHASE shift than series II (0.015 uh/foot is reduced to 0.11 uh/foot nominal). Lowering inductance directly lowers phase. See the QED phase analysis measurements on a variety of cable;

QED – The Sound of Science

OVERALL 4 x 1 wire XLR CMRR INTERNAL (signal energy)

The two MINUS fields cancel between themselves.

The two PLUS fields cancel between themselves.

But a MINUS to PLUS field REINFORCES the overall magnetic field.

The reinforcement makes the field stronger and the loop area effect worse.

BODY –To make improvements, we need to reduce the signal electromagnetic field to ZERO, in theory, both from an external interference view AND an internal electromagnetic conductor view. To do this, we need to BALANCE the music signal by SPLITTING each of the four SIGNAL wires into FOUR, or sixteen separate wires.

Making this critical change will theoretically remove the signal field currents that interact with the loop, creating inductance. It will also significantly improve the dielectric group and Phase delay by forcing the dielectric to be seen more uniformly across the 20-20KHz frequency range with smaller wires.

To keep capacitance low for a given loop area, we need to use AIR around the wires, and to make sure any plastics that touch the wire are super low dielectric constant materials (FEP mini X-filler and external FEP bead wire).  This is why the wires have to be BARE copper with NO insulation around them.  Only the tangential surface of the FEP filler and FEP beading, the rest is air. Capacitance is dielectric AND distance related where Inductance is distance and electromagnetic field strength.

Each of the four wire will be shorted together to make the typical four wires in a star quad. The wires are 10-mil diameter 30 AWG for a total CMA of 4 x 102 = 400 CMA. I used 4 x 0.018” in iconoclast for a total 18 x 18 = 324 CMA for each signal wire. 400 CMA is slightly lower DCR than the current design improving attenuation and mitigated ground loop voltages.

The collateral filler is foam FEP to manage capacitance. The power carrying signal braid should also be as far away as possible from the internal signal wire QUAD structure to lower the ground plane inside the cable, lowering capacitance. This means making the outer belting thickness under the braid to the best fit for an XLR connector, but not too big as the reduction in capacitance is a squared law variable, once a threshold is reached, more is not too beneficial.

The X-filler is FEP, as would the 30-mil beading wrapped around the QUAD wire to lower the dielectric nearest the wire where it is most critical. The material issues all control capacitance, not inductance.

The overall belt is solid FEP, with a 36 AWG BC braid and a drain wire. A final solid FEP jacket finished the cable.

CORE                    0.230”
BELTING             0.030”
BRAID                  0.015”
JACKET                0.030”
TOTAL                  0.305”

Does it really work on initial Capacitance and Inductance measurements? The final design using the ICONOCLAST™ all FEP design for ultimate performance appraisals measured as follows:


Lab Rqst-177575
Sample ID – 60156Y (PDC2842)

Requestor – Galen Gareis
Report Generation Date – 22 June 2017

Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Meas:  18.23 pF/ft

Inductance @ 1 kHz per ELP 424, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Meas:  0.10 µH/ft


Lab Rqst – 177587
Sample ID – PDC2842

Requestor – Galen Gareis
Report Generation Date – 29 June 2017

Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Cap @ 1 kHz Spec:  10.5 pF/ft max

Meas:  17.48 pF/ft

Inductance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Meas:  0.10 µH/ft

Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed.

Meas:  85.3%

4×4 Design1×4 Ref. Design
231 pF/12.67′ = 18.23pF/ft12.5 pF/ft
1.29 μH/12.67′ = 0.10 μH/ft0.15 μH/ft

To really get better XLR performance, both loop area and the field cancellation technology need to be leveraged, with the latter being most critical. The capacitance is all about materials and DISTANCE between them. Improving inductive field cancellation has the added, and significant, benefit of improving signal coherence through four smaller wires and phase with lower inductance while improving attenuation performance.

A cable with smaller signal wires and better coherence, low inductance (better phase) and slightly higher capacitance will sound better sounding than a cable with larger signal wire and less coherence, higher inductance and lower capacitance…as long as capacitance isn’t too high!

The prototype run does indeed lower inductance with the expected rise in capacitance.

0.15 uH/foot (X) + X = 0.1 uH/foot, X= 33% lower.
Or the current design is 50% higher.

12.5 pF/foot (X) + X = 18.23 pF/foot = 45.8% higher
Or the current design is 31.4% lower.

Since the original design is working from such low L and C numbers, the percentages are not really illustrating the advantages of the improved signal coherence with much smaller wires, and an advantage that should play out in audible performance.  The -3dB first order filter frequency is still well above the audio band so first order filter phase distortion is not going to be an issue. What must be the major contributor is coherence with the smaller wires. Rs response, while lower, is hard to quantify.

Rs (swept frequency resistance) Values

The 4×4 XLR lowers swept Rs (proximity effect) values significantly, and flattens the high-end linearity. Can you HEAR that improvement, over the single the wire design? The truth is BOTH are superimposed when the wire is used, and pushing the XLR designs to as near perfection is certainly a better and better design. The lower DCR is evident in the trace compared to the 1×4 25 AWG wire as is the flatter upper frequency measurements.

The RCA interconnect has also been updated with the new 1×4 (ONE wire made with FOUR conductors) design. The reactive variables will track with frequency like the single wire designs, but map to the altered L and C values.

The following table shows the effects of changing the wire size and number. The 4 x 4 has almost the same CMA as a single 22 AWG, but 1.82 times more total circumference, which shows up only at increased frequencies. The lowest frequencies are essentially DCR.

The maximum Rs is lower with the 4×4 design. Beta test feedback from customers on the 4×4 has been extremely positive and, consistent with the numbers, shows this revision to be a significant upgrade from the original Iconoclast design for analog applications.


The interconnect cables of  a given wire design (single to single and verses quad to quad) have essentially the same loop DCR values.

From the Rs chart above at DC, we see;

4×1 XLR and 1×1 RCA are 34.11 and 39.19 Milli-ohms/foot respectively.

4×4 XLR and 1×4 RCA are 27.53 and 28.79 Milli-ohms/foot respectively.

How was this done? The double braid on the RCA was necessary to mitigate ground loop DCR variation between sources, and the DCR was designed to be near a “free” return path for loop[ DCR. The loop resistance ios the braid plus the  conductor. But, the braid DCR is so low that the loop DCR is pretty much the RCA center conductor. This is true for eother design.

The XLR DOUBLES the number of conductors in each leg as a star quad. This reduces the DCR to one-half the conductor’s value. Thus the two pairs in parallel are the same DCR as a single conductor.

This was also done on purpose to make sure that the RCA’s loop performance was as good as the XLR, and that the RCA BRAID was essentially a ZERO DCR return path between grounds. If the RCA braid was insufficient DCR, we would see more divergence between the two singel ended and balanced design.


The measured XLR electricals are very good, and follow design theory perfectly. ICONOCLAST once again shows that proper engineering fundamentals are paramount to performance. Sound Design Creates Sound Performance!

XLR Design Brief

1.0 Conductors
1.1 Copper Size
2.0 Dielectric Materials
3.0 Dielectric geometry
4.0 Shield Material and design considerations
5.0 Jacket design and material considerations

1.0 Conductors.

  1. Copper Size.

BOTH of the copper conductor and size considerations were answered when we started the RCA cable. We don’t want to change the current coherence with a differing conductor diameter if we are to mirror the reactive variables, too. We need the same exact wire to shield reactive L and C parameters in each cable in the end configuration design. The geometry of each cable is entirely different so how to you do that? That is, assuming you want to match the RCA and XLR properties and maintain the same signal quality…and we certainly do.  There is no reason to copy a bad sounding RCA cable when designing an XLR, so the RCA is designed FIRST.

  • Dielectric material(s).

One difference in the XLR is that we are going to use FOUR wires in a star quad configuration. (Note: in our “Gen 2” XLR product, there are sixteen wires — four wires making a star quad in place of each single wire in the design shown below.  For more detail, see the last paper in this series.)  Four wire XLR cables use two cross-connected wires for each polarity, which doubles-up the wire AWG for lower attenuation. Two 25 AWG have the DCR of a single 22 AWG yet has way better signal coherence by using smaller wire.

I could have used a cheaper and easier two wire XLR design but the inductive and signal coherence benefits of a star quad are too good to pass-up. If I can get the materials and quad design to achieve a high enough level of performance it is a better cable design.

Star quads have a higher degree of CMRR (Common Mode Rejection Ratio) when properly signal balanced. There are three primary reasons for this;

  • The two or four wire stranding “twist”.
  • The differential encoding.
  • The outer shield properties, but only at RF frequencies.

Two wires of a star quad are a “positive” voltage, and two wires are a “negative” voltage (180 degrees out of phase), hence the term “balanced”.  If the cable were a teeter-totter, it would sit level. Some call this differential mode since each signal is equal but different.

Differential Mode Transmission
Perfect Wire Balance Equals Less Noise

In the example above we show two wires, but the system is the same in a star quad. The signal we WANT is encoded as +2 volts and -2 volts. The noise can’t “change its spots” relative the cable’s twisted pairs and shows up as the same voltage on each wire, +1V noise in this example. The TWIST ratio helps make sure that the wires see the noise the same amount of time and this is vital to the function of the circuit.

Here is where the balance is so important; the signal IDEALLY becomes the superposition of ALL the voltages, or +3 volts and -1 volt. No more, no less. The signal voltages are STILL exactly 4 volts “apart” from each other; +2 to -2 with no noise and +3 to -1 volts with the noise.  The signals are fed into a difference amplifier that, you guessed it, looks at the “difference” between the two voltages and see’s 4 volts with, or without, the noise. The noise is absent in a perfect world at the difference amplifier’s output.

In order to do this, every wire has to be presented to the noise in the exact same way via the cable twist and has to be the same length so the signal stays TIME aligned down the wire and has to have the same attenuation. The difference amplifiers need to be nulled perfectly between gain halves. Believe it or not, this gets done really well with good quality products.

The control tolerance of the copper is 0.0005”, so attenuation issues are mitigated and CUB (Capacitance UnBalance) tests insure we see MIL standard quality in the finished cable. All quality types of copper can be used in the XLR design. It is the overall structure that is the most “magic” and not as much the copper itself, although the copper draw process does influence the sound.

We have several variables that aren’t present in a coaxial cable design to contend with;

  • CUB, Capacitance Unbalance or, each wire shows a differing capacitance to ground.
  • DCR unbalance, each wire has to be the same DCR.
  • CMRR remainder, the differential signs have to NULL to the exact same point neither above nor below reference ground.
  1. Dielectric geometry.

Lots of words, time for a picture;


The above CAD drawing is what we have inside our XLR design so far (well, I ignored two wires in the drawing).

Remember I wanted to make L and C reactive variables EXACTLY the same for each cable with EXACTLY the same wire size and draw science? What else do we know? I also said that CAPACITANCE is sensitive to the distance to a conductive plate area, and that means ALL the way around the wire. The coaxial cable is easy; we purposefully put a ground around the wire at a known distance that defines the capacitance ground plane reference distance and inductive loop area.

In the coaxial cable, the center of the wire to the inside of the tube is 0.098” / 2 = .049”.  Ok, so what? This is what. The capacitance is a squared law property and predominantly sees the ground closest to the wire. The shield on the opposite side of the XLR cable, to a first approximation, falls a way. We actually measure the capacitance BETWEEN the two cross paired wires but the ground location still influences the capacitance. Also, we have four wires that are capacitors.

This doesn’t “sound” good, does it? We have four times as many wires and all have capacitance. Somehow this is supposed to come out around 12 pF/foot (with connectors), same as the RCA!

Now for the inductance part, L. Inductance is loop area defined. It could care less about the dielectric, but the graph above shows a HUGE ~0.170” loop area! How is THAT going to get to the 0.15 uH/foot inductance of the coaxial cable? I could make something up, but that isn’t as neat as what’s really going on.

To get capacitance as low as I need it to be to match the coaxial cables, I use DISTANCE between the wires. And yes, this DIRECTLY sets what the inductance will do…hold on a minute.  By using AIR, I can set the C-C of the wires to meet my capacitance target needed for the final tested value with two cross wires connected and tested between them. AIR lessens this distance for a given value of C so I can also manage inductance now. For inductance, L, the smaller the wire loop area the better for a given value of total capacitance. Air gets me far closer than any other dielectric.

How much air? Well, EXACTLY the same as the coaxial cables! How do we do that? The standard answer is, “very carefully”. Let’s look at a drawing;


Still, so what? Yep, I agree, until we compare this area to the area in in the RCA cable air dielectric; 0.00754 in2. OK it isn’t exact; I missed by ~0.000009” in2.  I use the exact same thread design around each identical wire so it’s all the same area in the chamber as in the RCA.

Let’s do some reality checking as to what it SHOULD be based on MEASUREMENTS and calculations.

  • We have the EXACT (can I say that as close as it is?) same velocity of propagation based on the composite (air and plastic inside the ground plane) dielectric; 87% at RF reference.
  • I measured the IMPEDANCE at RF @ 100 ohms, same as the coaxial cable.
  • The dielectric constant can be calculated and from that the VP, VP = 1/ SQRT (E).
  •  And from that composite dielectric I also know what the capacitance has to be.

Capacitance (remember that chart on dielectric value and capacitance earlier?) is directly linked to the group dielectric constant. I know VP, and I know the impedance, so I can calculate the capacitance and then get the dielectric constant from that.

101670 / (C * 87) = 100 ohm

C = 11.68 pF/foot.

What does the cable actually measure on capacitance? The chart below shows 11.767 pF/foot. Notice that the capacitance values between each of any two wires has to be ~ 5 pF/foot to “double-up” the two wires capacitance and still to arrive at a final ~11 pF/foot! Yep, that’s LOW capacitance. Capacitance adds in parallel so this is a significant issue when a design uses four wires.

Below is the measured and calculated imbalance of the capacitance between 1-3 and 2-4 cross wires’ conductors as a “pair”; 2.02% unbalance, very low.

We seem to have the capacitance and VP looking much like the coaxial cable. Remember, measurements include ALL the approximations in the soup.

So what about inductance with that WAY larger loop area? Isn’t that going to really kill this thing?  No, because of some properties of magnetic fields. Magnetic fields CANCEL if they see each other in OPPOSITE directions. Inductance is the “reactance” or “resistance” to instantaneous flow of current. If we can REDUCE the magnetic field lines, we can directly reduce the measured inductance.

We also know from the basic equations that DISTANCE between the two wires is important. Keeping BOTH distance and magnetic field line magnitude small lowers inductance, and removes the noise.

The picture below shows what’s going on…sort of. For now, we’ll pretend the field’s ONLY go “inwards”, or inside the wire, and stop there (they don’t). If the lines that extend outside each wire do the OPPOSITE as the field INSIDE the wires, they reinforce the field!  It is generally accepted that the flux lines concentrate substantially BETWEEN the wires.

If we draw arrows that represent the DIRECTION of the circumferential magnetic field waves AROUND each wire we get what is shown below for a NOISE signal hitting the wire. We have TWO different voltage polarities so we have TWO different current directions for the SIGNAL, but the NOISE is the SAME direction in all the wires.

If you grasp a wire with all four of your fingers, and point your right hand THUMB in the CURRENT direction, your fingers will point in the field’s circumferential direction around each wire. The arrows are a “part” of the noise current field lines “inside” the four-wire group.

NOISE FIELDS (all the same direction)

Where the arrows are OPPOSITE each other in direction between any two wires, the field lines cancel. For NOISE every field theoretically cancels. ADJACENT or ACROSS from any TWO wires we induce field cancellation with a star quad design.

For the SIGNAL, we now have TWO equal but opposite current directions.

This allows larger wire-to-wire spacing in order to lower capacitance and also keeps inductance low. Inductance is managed with field line cancellation geometry.

Now we know why I didn’t use a two-wire system, you can’t manage CMRR.

Let’s look at the situation for the signal. Below is a simple picture of the field cancellation between four wires with opposite polarities wired as a star quad.

SIGNAL FIELDS (opposite directions)

   Minus = Current INTO the page (CW rotation)

   Plus = Current OUT of the page (CCW rotation)          

Reduce Signal Loop Area to Reduce Inductance

What do we see? The all the signal field lines DO NOT cancel. Adjacent wires reinforce, and opposites wires cancel. Reducing loop area is the best way to manage inductance because we can’t cancel all of the field lines, only some of them. This theoretical field relationship limits the ability to reduce capacitance for a given inductance. Using low dielectric constant materials to lower capacitance (Air!) allows closer spacing needed for low inductance.

Is there a design that can, in theory do BOTH, reduce signal and noise fields to “zero”? Within the limits of DESIGN, yes there is. The ICONOCLAST series II reduces both noise and signal field cancellation. The wires, in practice aren’t EXCATLY the same distance apart and EXACTLY the same resistance, so we say “in theory”. But, reducing the nose to 1000 or more times less and reducing the inductance 27% is indeed achievable.

So, after all that explaining, how does the star quad ICONOCLAST cable measure up? Tests at 1 KHz show the following values below. The inductance between the two cross wire pairs of the star quad are 0.15 uH/foot inductance…same as the RCA.

So what does the “reactive” picture look like comparing the RCA and XLR? How close are they to being the same? This swept test is the real deal. There are no approximations to fudge.

What we see above is impedance / phase for the XLR and RCA superimposed one on top the other. Note that there are four separate lines. We have two identical cables with exceptionally low reactive variables.

  • Shield material and design considerations.

There is yet one last thing to consider in the XLR design; the outer shield. A 95% BC (Bare Copper) braid is used. Audio cables are not RF designs, and the braid shield will NOT shield low frequency magnetic interference. The CMRR of the XLR is going to do that for us. Excellent CUB, DCR unbalance and twist ratio all aid CMRR. The braid DOES knock down RFI by 80 dB, so that’s a given. The shield isolation @ RF mitigates NULL balance at high frequencies only.

Like it or not, 20-20K is a predominantly magnetic field frequency range where the B-fields decay at a ratio of 1/x^3. DISTANCE is the best solution for isolation of cables with magnetic properties.

5.0 Jacket design and material considerations.

All ICONOCLAST cables use FEP as the jacket to reduce UV sensitivity, plasticizer migration and provide chemical resistance.  The cables are designed to last decades.

I hope that this design summary of ICONOCLAST RCA and XLR interconnect cables shows how important good design is for ALL your audio cables, and that every manufacturer has to manage all the same variables to produce these results. There is little “magic” in the design of good cables. There are indeed tertiary variables that we can’t measure, but those should not influence the ones we can measure, or at least not excessively so. Mother Nature abhors complexity, so the better managed the known variables in a cable are, the more properly it may highlight “unknowns.”  To put it another way, the more we put knowns into their proper place, the better we may distinguish the effects of the unknown. Wire draw science, for instance, can be heard better, and more fairly, in a superior electromagnetic design.

Belden appreciates your interest in how quality interconnects are made, and how / why ICONOCLAST RCA and XLR cables were physically derived as you see them in their production form. We have no special sauce or magic in our products, and I think that the cables perform as well as they do BECAUSE we did not design around “unknowns” and then make it appear as though we had unique influence on those unknowns in the design.

Truly low R, L and C cables are difficult to make when consideration is given to all three variables to manage them in a truly balanced fashion. The designs can be frustratingly simple looking but hard to manufacture, as processes are pushed to the limits of current capabilities. Belden’s focus is to make real measured values as low, and properly balanced, as we can. ICONOCLAST interconnects represent the pinnacle of low frequency measurements and electrical balance between the RCA and XLR (same electromagnetic properties).

The next design analysis will look at the SPEAKER cable.

RCA Design Brief

In a previous paper I covered several issues that create signal distortion in audio cables. The most demanding variables involve the TIME related distortions that the ear is most sensitive to. Consideration must be made during cable design to mitigate the TIME based issues through the audio band. The following paper is the journey through the design process to arrive at a satisfactory RCA and XLR cable design. I must stress ALL quality cable designers have to work with the exact same known variables to solve problems at audio. Every cable is a compromise of some sort as distortions can’t be eliminated.  ICONOCLAST has made the outlined design decisions to arrive at, what we think, is an industry leading design based on real measurements. 



1.0 Conductors
1.1 Copper Size
2.0 Dielectric Material(s)
3.0 Dielectric geometry
4.0 Shield material and design considerations
5.0 Jacket design and material considerations

The design process will start with the RCA cable as this provides the most pristine electromagnetic properties possible due to the seemingly simplistic design.  Once all is said and done it is “simple” looking. The more complex XLR will have to, somehow, match the RCA’s electromagnetic properties if it is to be an “equal” on measured attributes. If the RCA isn’t any good, I may as well start over again!

  1. Conductors / RCA

There is a lot of mystery around copper. The grains, the molecular arrangement of the crystals themselves were recently found to NOT be what we thought;

“…granular building blocks in copper can never fit together perfectly, but are rotated causing an unexpected level of misalignment and surface roughness. This behavior, which was previously undetected, applies to many materials beyond copper and will have important implications for how materials are used and designed in the future…”

The battle for material supremacy continues. However, what we tend to discount is that while the overall design of the tire we put on the car is important, the rest of the car has more to do with what that tire does than just the tire. We over spec the tire and vastly under spec the car. I’m intent on building the car, not the tire.

The decision to use copper is based on several factors, none of which were price. Copper offers the best material for affordable cables with a significant level of performance in more ideal electromagnetic designs. Far more expensive materials in lesser designs won’t work, and far more expensive materials in superior designs won’t work…for most of us anyway.

Copper is available in several process treatments and after process treatments;

ETPC (as good as what used to be OF grade)

OFE (differing process, but far from vastly lower impurities content)

UP OCC (what is often called long grain type, and again a differing process).

Cryo treatments (used to improve copper’s PHYSICAL properties)

Grain direction (music is AC. Which polarity do you like first and at what frequency?)

I don’t use wire “quality factor” as a design element since every contemporary draw science wire is of vastly better quality than ever. Sure, some processes are more $$$ but there is scant repeatable measurement that I can do other than conductivity, a passive resistive measure that will influence R, L and C. The conductor type is an option for the customer to listen to, only. There are differences. Belden just isn’t in the position to create a pet project to define what isn’t yet scientifically defined. That’s not our thing.

Belden offers the three fundamental copper grades; ETPC, OF and UP OCC, as they DO sound different in the exact same electromagnetic R, L and C referenced design. No changes other than the copper, so we know what the culprit is. What we don’t know, is WHY it is the culprit. Instead of making up a big old story, again, about the material, we don’t. It is what it is in use and we leave it that way.

What we don’t offer is what I can’t hear as a designer. Sorry, but I’ve yet to hear CRYO treatments, intended to improve the wire’s PHYSICAL strength or grain direction, change the sound. As far as grain direction goes, you can flip the leads in any direction you want, as the wire’s grains all go the same way due to the manufacturing process that we use. If you can hear the direction switch, flip them any way you like. We won’t send you a bill for that!

Any material used in a superior design SHOULD sound as good as it can, and cost isn’t a direct line to better sound. I ignored cost when I designed ICONOCLAST™, either high or low. If my system didn’t allow me to hear it, I didn’t use it (materials) or do it (process / design).

This isn’t a paper on conductors, although I may have some things to say about alternatives to copper them later on in another paper based on some measurements and calculations I’ve done. We’re talking copper in this paper as it is the very best economical solution that we have right now.

Copper has a very low DCR, a reasonably deep skin depth to manage current coherence, is pretty high in tensile strength for processing, and in most applications resists severe oxidation. The grain structure is clearly visible in form, but that alone is NOT what makes the different grades sound different. It is a trait of the draw science, but does not have as much effect on the  sound as you would be lead to believe.

Use solid or stranded wire?  This, at least, is easy. Is stranded better for the way the cable is used? Is stranded more, or less, expensive? Is stranded easier or harder to process? Is the termination of the cable better or worse with stranded wire versus solid? Are any gremlins that I call tertiary variables (stuff there isn’t a measurement or calculation for) removed if the truly measureable variables are accounted for between stranded and solid?

ANSWER – Solid wire wins hands down for this application. Every question is in solid wire’s court. End use, costs less, processing cost, ease of termination and lack of tertiary elements (all those diode effect “arguments” between strands and more).

On that, though, a note: the first generation of Iconoclast interconnects use single solid wires for the signal-carrying conductors and that’s what’s discussed in this paper.  Our second generation product (suitable for analog but not for digital due to impedance issues) uses a star-quad arrangement of four separate wires, placed around a separator, in place of each of these conductors for improved inductance; for details see the fourth paper in this series.  Other than this change in the signal conductors, the “Gen 1” and “Gen 2” interconnects are the same.

1.1       Copper Size / RCA

We now have SOLID copper wire. The size selected sets the foundation for the whole thing if we consider that the cable’s structure is supposed to allow a conductor to be as near zero R, L and C measurement cable as we can design. 

You can’t use a conductor you can’t process. For the RCA cable, we want as small a wire as we can process as this will force the best current coherence through the wire (same current magnitude at all frequencies). The exact skin depth calculation is a tool we use to gain the knowledge to reduce the wire size in audio cables. At RF, we use it to tell us how much copper to put over a STEEL support structure to maximize RF attenuation. Audio is not RF, and the ENTIRE wire is used to move the signal and at ALL frequencies concurrently, not the same issue at all in RF cable design.

RCA cables terminate into a theoretically infinite (47K-120K or there about) input resistor. We say impedance, but it is really as resistive as it can be made at the input op-amp level. Yes, purists will point out that input impedance DROPS some at higher frequencies.

If the impedance is so high and the current is so low (it looks like an open circuit) just use as small a wire as you can! Well, yes and no. It has to be reliably terminated and secure in the end product, and it has to process evenly under tension and not fracture from surface issues.

A review of the end of process design backs into the initial design requirement. Calculations and testing selected a 0.0176” diameter wire for ICONOCLAST. The process has to handle less than 4-3/8 pound tension to avoid permanent wire stretching. Wire was tested for the process requirement.

The 0.0176” diameter wire (0.0088” radius) is one half the diameters necessary for one full 18-mil skin depth at audio, so we have significantly improved current coherence through the wire @ 0.0176” diameter wire. Skin depth is FREQUENCY driven for a given material. The smaller the wire the larger the inner current magnitude will be relative to the surface current. We want as good a shot of that as we can get.

The RCA cable’s loop DCR will essentially be the center conductor in an RCA, if it is made right, and ICONOCLAST is. The center wire governs attenuation. The outer conductor is, in theory, infinitely low impedance so it nearly drops out of the loop DCR calculation and leaves the center wire.  The length of the cable relative to the input impedance allows a SMALL wire at audio. At least attenuation works in our favor at audio as it is a LOG relationship and gets really high very quickly as you go up in frequency. For audio, we can relax a bit on attenuation as it is low for the lengths we use and is in the right frequency range to stay low. Attenuation is a passive “distortion” and is VERY hard to hear over TIME based distortions.

  • Dielectric Material(s)

We’ve already made a critical choice in our cable. The wire material and size. We’ve used good engineering practice to KNOW what the decision will yield. Now, how to RETAIN all that the material / size wire can provide? That’s easy, just stick it in air and find an infinitely low ground potential for our unbalanced / single ended wire!

OK, this IDEA is easy. The execution isn’t. I don’t care about speed of the process and / or costs as I’ve used REASONABLY affordable material as my conductor. We can always go back and break the bank on conductor materials. AIR is free, but expensive to get. Air is by far the best dielectric to have, and especially nearest the wire were the influences are the worst on group delay. The closer to the wire the dielectric is, the more it influences the overall velocity of the composite structure (wire / beading/ then plastic tube thickness / then braid)

I decided to go the tough route and use air. We can use RF as a HINT at what to do overall. We have used designs called semi-solid core dielectric RF cables. These partially suspend a wire in a tube with a spirally wrapped thread. The problem is that the wire SIZE and the core tube properties aren’t suitable for audio frequencies. Even the choice of materials isn’t as important at RF as we can reach a set impedance vector (real + the reactive inductive and capacitive parts all added together) by tweaking the thread and tube dimensions.

3.0 Dielectric geometry

The audio signal is very sensitive to the dielectric effects of the plastics near it. I chose a specially made beading thread to get the job done.

The above picture beading around the wire is a glass thread coated in pure TEFLON®.  I use a ROUND beading shape versus square, as it touches the wire at the tangent points for the very LEAST effect nearest the wire. The electromagnetic field sees the entire cross section of the plastics and material between the wire and the inner braid, so I use GLASS thread inside the beading as it is a good dielectric, too. Why is the glass there? A solid TEFLON® bead can’t be processes at this size and keep consistent dimensional linearity. The glass is the true STRENGTH member in the beading, not the plastic. The plastic is to set and hold the shape. The glass lets me process the beading at production speeds.

Why TEFLON®, really? OK, I’ll tell you. It has the lowest dielectric constant of any solid plastic. It is TOUGH in thin walls for end product dynamic stability; the bead should STAY round under side-wall pressure. This is a SMALL bead, so I need that toughness. TEFLON® has high T and E’s (tensile and elongation) properties for process toughness. We don’t have much process room, as I’ve calculated backwards how big this bead would need to be in this design and wire size.

How big should the conductor be based on a tube ID? There is ONLY one optimum asymptotic wire size driven MAX AIR volume (%) based on the tube ID. The ratio of the tube ID with the 80% air void to the inner braid surface will determine the capacitance. Maximizing the air content will improve the efficiency of the dielectric so the smallest loop area for inductance will also yield the smallest measured capacitance.

Here is what happens when we CHANGE the wire size;

Tube ID (IN)Wire Size (IN)Air %

As the wire gets bigger or smaller inside a given tube ID, it crowds out the air. We COULD go drastically big in the ID of the tube and wire size (0.150” tube ID)…but we want to hold INDUCTANCE and signal coherence in check. Inductance is the loop area between the wire and the inner braid, and that needs to be infinitely close, the opposite of capacitance. For a given tube ID size we want the maximum amount of air void and the smallest possible wire to braid distance. This means the conductor wire size has to be as small as you can process, and with the desired capacitance. As the tube ID gets larger, cap will drop but inductance will rise, and the opposite with a smaller tube ID. The design target is 11.5 pF/foot on the bulk cable to assembly capacitance would be 12.5 pF/foot.

Using too large a wire hurts frequency coherence so we pushed the wire size DOWN until inductance was moving off spec relative to capacitance. A balance was sought between wire size (coherence) and reactive variables (L and C).

I can do a quick check to see how I’m doing by applying a test ground over a ten foot sample. Using RF frequencies as a “constant” since the velocity has stabilized to an asymptotic maximum, we measure really high VP values, ~ 87%. This is good as it allows me to reference to end capacitance, too. I just treat the cable like an RF cable and work the capacitance backwards from the open – short Impedance; Z = 101670 / Cap * VP. This is about 104.6-ohms so capacitance calculates to 11.2 pF/foot versus a measured value of 11.19 pF/foot.

We know from the previous paper that Capacitance and Inductance are FLAT with frequency, and are actually measured at 1 KHz. Our 11.19 pF/foot bulk cable value is true at 20Hz-20KHz. Inductance is a low 0.15 uH/foot through the audio band as well.

Capacitance @ 1 MHz per ELP 423, Agilent E4980A Precision LCR Meter, Belden’s Cap/Ind Test Fixture

     Spec for Cap @ 1 MHz: 12.5 +/- 1 pF/ft

     PDB1610 B24 Cap @ 1 MHz: 11.1947 pF/ft

Characteristic Impedance per MIL-DTL-17H (ELP 142) using the included equation:

Char. Imp per ELP 142:  Imp =    101670/(C +VP)

       Spec for Impedance: 100 +/- 5 Ohms

       PDB1610 B24 Impedance: 104.631 Ohms

       SEMI-SOLID PDB1610 finished RCA “assembly”

CAP                12.25 pf/foot
IND                0.1450 μH/foot

Inductance isn’t as critical in high impedance leads as current, which is ride time limited by inductive reactance, which is near ZERO, but in my listening test, cable with near zero on BOTH L and C attributes sounded best, and a BALANCE needs to be considered. The cable isn’t big or small; it is what it needs to be to WORK. The wire size we start with sets this all into motion.

The FEP tube is critical to get right. Special processes are used to keep it on-sized and ROUND over the beaded center wire.

  • Shield material and design considerations.

We have a core tube and know the electricals, so now what? The braid is much more important than people think, and for a different reason than people think. No, it isn’t shielding, either. True, a double 90%+ braid have 90 dB RF shield properties but, I sure hope your equipment isn’t THAT sensitive to RF. Foils are much better and more economical for RF than a single 80% braid and the shield reaches the 90 dB mark far more cheaply.

RF cables are “shielded” to RF noise and IMMUNE to low frequency nose (outside their pass band) because the shields have a low resistance to RF, measured as transfer impedance. This is sort of like low DCR at audio frequencies, but relates to how high frequencies work. Audio cables are not RF cables!

We need to look at how unbalanced circuits work. They SHARE a ground…or do they? They are SUPPOSED to SHARE a ground. They don’t. RCA unbalanced cables use the CHASSIS as a ground to the wall outlet or it is floating in some cases but the REFERENCE between the grounds is still there. In ALL cases, there is that pesky WIRE thing called the SHIELD between the ground points on every piece of RCA equipment you use. That wire has RESISTANCE and that resistance creates a ground potential difference so current starts to flow between the two end grounds. E=I*R, remember that? A VOLTAGE is impressed against the center wire and the magnitude of that voltage is the current times the resistance. We can CONTROL the “R” by using TWO 98% copper braids. This is $$$ to do, but it is the RIGHT thing to do.

No, those braids won’t shield MAGNETIC interference. The HUM you hear is more than likely ground loop current through the braids resistance called SIN; Shield Induced Noise. The lower the braid DCR is the better the SIN rejection. You need low permeability shield to block low frequency magnetic waves (anything below about 1 MHz starts to have a considerable B-field bent over E-field). Good audio RCA cables ARE NOT going to shield B-fields. They will shield E-fields and reduce SIN noise.

To shield magnetic B-fields a MAGNET needs to be able to STICK to the shield. This is an indicator that the material is “influencing” the magnetic field flux lines INTO the metal and OUT OF the air. We can manage the SIN noise with a good ground, but true extraneous magnetic noise is still tough with unbalanced cables. Now you know why. It’s the ground system it uses. 

  • Jacket design and material considerations

ICONOCLAST uses an FEP jacket for some good reasons. FEP is the most chemically inert material there is, protecting your cables from chemicals and UV exposure through those nice picture windows in your house. Lesser plastic material isn’t as stable, or inherently flame retardant. Nor can many materials be used in thinner walls.

Plasticizer migration out of the cable, especially near heat, is a real issue in contact with polyester or nylon carpet that would love to be the same color as your cable laying on it! My previous cables were.  FEP does not have this issue and will look nice for decades to come. Yes, it costs some more but these cables are an investment into the future and can follow your system several steps above where you may be now. Based on durability, stability and inertness to solvents, FEP is the best choice for the long haul.

RCA SUMMARY – Knowing that RCA cables aren’t as “shielded” at audio as we think, what can we do about that? If you don’t have the problem, you’re good to go! RCA is a great sounding cable by fundamental electromagnetic design. This is why it was created. It does have magnetic noise immunity issues, though. There is no magic to good cables; it is adherence to strict design rules that also encompass those “magic” tertiary variables called wire science.  The same design adjusted for a new material’s skin depth properties can be made to the same “ratio” and match the electricals with differing wire.  The layers of the onion and their thickness can be altered (L and C values) depending on what is most audible. Tests won’t tell you that, this comes from design experience.  This does NOT mean that either L or C can be thrown to the wind.  Both L and C cause TIME based distortions and neither is welcome in good cable.

Then there is the next cable I’m going to talk about that does exactly that, except it is far, far harder to make as good as an RCA electrically. It is called the XLR cable.


If you have spent plenty on cables you may well wonder WHY these cables are physically as they are. If care is taken to adhere to fundamentals, there are very good reasons for a physical design in audio cable, of both high (interconnect) and low (speaker) input impedance types. If we look at all the fundamental electricals through the audio band, is it any wonder every cable doesn’t sound different? Let’s see why that might be, and no magic need apply throughout this analysis.

What is happening in audio frequency ranges?

  1. What exactly are we “moving” with zero distortion?
  2. Current and Phase Relationships.

3.0 Electromagnetic wave propagation differences with respect to frequency.

4.0 Impedance and matching to a load at audio.

5.0 Capacitance and Inductance with respect to frequency.

6.0 Cable Capacitive and Inductive reactance properties rise and decay time distortions.

7.0 Current normalization / skin effect.

8.0 Dielectric effects.

9.0 AC resistance changes and frequency.

10.0 Cable symmetry issues.

11.0 Attenuation at audio.

12.0 Passive low pass filter effects.

If we look at pure tones; sinewaves, square waves, frequency and TIME are interchangeable. Math says that this is so, and there isn’t anything new that explains that away. When we add TIME based distortion to the sound delivery system our ears are quick to “hear” the deterioration in fidelity based on frequency arrival time and phase coherence more than amplitude limitations (attenuation).  How much is a cable responsible for this?  The superposition of the 12 listed distortions (and there are more) are much more significant than any one taken on its own. There is truth to the concept that slew rates, or how fast a system responds (wider bandwidths), affect performance. A square wave is but a multiplicity of sine waves. Mathematically every frequency’s characteristics, at every point in a cable can be predicted.  Cable is far from perfect at moving electromagnetic wave through the audio band, however well we can calculate the accumulating TIME based distortion as the electromagnetic wave moves down the cable. Better designs minimize those distortions and place more or less emphasis on each one depending on the designer engineer’s concept of audible influences. The fact remains, cable design is still driven by the DESIGN needed to reach the R, L and C values with minimal influences on tertiary elements. Can you hear a more fully optimized design? This is why we present these designs for audition.


The issue – What do we actually LISTEN to on a cable? What is the “root” reason to be for a cable?

Cables exist to move the “signal” from one place to another, but few really consider WHAT that signal is. The signal we “use” is the electromagnetic wave moving down the cable at the group velocity of propagation of the dielectric. OK, what did I just say? Imagine our wire surrounded by a donut with a hole in the middle! The electromagnetic wave is this donut. There is an ELECTRIC (E-field) around our wire too, but this field is attached to the donut radially, and ninety degrees orthogonally to the donut’s circumference. To make the E-field, take a bunch of tooth picks and stick them all around the outside of the donut, that’s the E-field.

Now we have two imaginary waves, one low frequency and one high, sitting there. To MOVE that field, electron flow starts it happening.  To keep it simple let’s distort our wire to be a TUBE full of marbles (electrons) that has an inside diameter the same as the marble’s diameter. To make the magnetic field move, and drag along the E-field with it, we apply an electromotive force (electrons / marbles) to the tube. When a marble is inserted into the end of the tube, the marble at the opposite end pops out as fast as the marble can be inserted into the send end of the tube. This “speed” is determined by the velocity of propagation of the dielectric, or the tube in our case. Something funny happens with the magnetic field though; it follows the PROGRESSION of the electron (marble) flow. When the marble is half way into the send end of the tube, our donut with all our toothpicks (the B and E fields) is halfway down the cable already!  When the marble is inserted all the way in at the send end, the B and E fields are at the END of the cable. So the “signal” we use travels at the VP (velocity of propagation) of the cable, and NOT the speed of the electrons at all. Those move very slowly compared to the electromagnetic B and E fields. Now we have the donut at the end of the cable. But, we won’t ever see a baker’s dozen, or zillions more moving electrons appear at the same time at the opposite end of the cable if we carry more than one frequency concurrently since every frequency has a different VP through the audio band.  All individual frequencies will have significant arrival time “distortion” between frequencies. In other words, every marble that represents a frequency in my example is inserted at a different speed (Velocity of Propagation) depending on the frequency the marble represents. Ideal cable should move a signal (now we know it is the B and E fields) down a wire at the same speed and shape at all frequencies. It doesn’t.  

2.0 Voltage and Current Phase

The issue – Current and voltage are locked into a phase shifted relationship, always.

The reactive properties of inductance and capacitance are responsible for a ninety degree time based shift in all electronics, not just cable. There is a common ditty about the current to voltage phase relationship that goes like this; “ELI the ICE man”. It is a memory tool to remember that voltage (E) leads current (I) in an inductor (L) and that current (I) leads voltage (V) in a capacitor (C).

Why is this? A capacitor has to charge with applied current to reach a steady state voltage, so as the voltage potential increases the current drops. The current has to be there BEFORE the voltage potential hence current leads voltage in a capacitor.

An inductor resists current change when voltage is applied. Current reaches a steady state over TIME with applied voltage, so as the current potential increases the voltage drops. The voltage has to be there BEFORE the current potential hence voltage leads current in an inductor.

These two locked-in relationships lead to all sorts of other TIME based issues in cable and circuits. They are the variables that constitute PHASE in an impedance trace, for instance, and reactive TIME CONSTANTS that we’ll cover later in the paper.


The issue – VP varies the arrival time of signals moving down a cable. Signals should ideally leave and arrive at the same time and shape as they are sent at all frequencies.

Audio is in an electromagnetic transition band. This is the elephant in the room. It prevents cable from EVER being perfectly accurate when moving low frequency electromagnetic waves. The propagation constant, the speed at which the electromagnetic wave / signal moves down the wire’s outer circumference, and not IN the wire, is determined by the dielectric material that the electromagnetic wave is predominantly traveling through. We can measure this effect directly and indirectly.

At RF, where life is way more consistent for cables, we can calculate the velocity from the DELAY equation. For Ethernet cables the following equation is used;


The delay equation uses FREQUENCY. This is a TIME based value so it tells us that we have arrival time issues as the frequency changes, and less so at RF, and WAY more so at audio frequencies. The table illustrates the slow erosion of speed as we reduce the RF frequency. A little change is evident but audio frequencies see much more change.

Delay values measured at RF (MHz)

Actual data shows what audio cables do; the impedance RISES as we go LOWER in frequency, by a lot. This is because the DELAY / VP factor drops, and adds TIMING issues to signal delivery.

Above are actual traces of how ICONOCLAST performs across the audio frequency band vs. typical zip cord speaker wire (1313A) and out to RF, to prove a point. The impedance increases considerably below the RF frequency reference values. Those 87% and 90% VP factors we love to “hear”, high VP, are clearly not valid in the audio band.

How significant is the VP change? In the example above we drop from ~110,000,000 m/Sec @ 20 KHz to ~5,000,000 m/sec @ 20 Hz or a factor of 22 times slower through the audio band.

To make matters worse, it is a LOG function so it is not linear. This is what physics has thrown into the design process.  Can we hear this change? Attenuation at audio is a passive linear variable and considered to be insignificant (keep your cables short) but every variable keeps adding up to the overall actual performance.

Notice that the cable’s impedance, made for audio not RF, flattens out @ ~ 50 ohms above 100,000 Hz (see the table below for the actual values). Just because something has an “impedance” (real and reactive L and C component) does not mean it is a transmission line.

Look at the low-frequency range. Isn’t cable supposed to be the same at all frequencies or the same TIME base? The velocity constant at a frequency is TIME, so the fact that we see a difference indicates a non- linearity across the usable audio band. The problem is that thing called propagation velocity (VP) or the speed that information travels at differing frequencies in the cable.

The equation at audio compared to RF is more complex (wouldn’t you know it!);

Z = sqrt((R+j*2*pi*f*L)/(G+j*2*pi*f*C))

impedance (Z),

capacitance (C)

inductance (L)

resistance  (R )

conductance (G )

Using the general simplified RF equation, where all the extra stuff in the complicated impedance equation at audio goes to a one or a zero and drops out, we are left with; 101670 / (Capacitance x Velocity) = Impedance. At RF for ICONOCLAST speaker cable;

101670/(VP*45pF/ft) = 50 ohms @ RF

     Solving for velocity of propagation we see it is no higher than 45% at RF. This isn’t RF cable, and the design changes necessary for audio are what ICONOCLAST is after.  We need to ideally FLATTEN the VP curve for audio cables to better time align the signal in the frequency range where we use it.

The calculated graphs using a 75-ohm coaxial cable below show that VP change as we go lower and lower in frequency. Look at the IMPEDANCE at audio frequencies shoot way up, and the VP drop like a rock in a pond. Notice, too, that VP begins to flatten out at 100,000 Hz, just like the charts above on ICONOCLAST. This is real stuff, and it won’t go away…you have to MANAGE it to a balance in each cable.

What does our measured data show that corresponds to the theoretical chart above? Below we see several BELDEN products measured VP drop considerably from RF to, and through, the audio band. And, the measured values are near the exact same values I will calculate from measurement on ICONOCLAST; ~ 5% VP to 50% VP between 20Hz to 20 KHz.


The impedance goes up as we go lower in frequency because the velocity keeps going down, and the alternative variable, capacitance, just sits there (we’ll get to that soon). We have a differential in signal velocity across the audio band. Also notice that typical 1313A ZIP cord behaves much worse than ICONOCLAST™, rising to double the ICONOCLAST reference impedance value. Be warned, audio cable does NOT respond to impedance matching like RF.  

Speaker cables are theoretically designed to be much lower impedance, and terminate into reactive 2-16 ohm loads, and some point way north of 16 ohms.  Interconnect cable is terminated into “high” impedance resistive loads of 47K to 120K or higher, and should be much higher theoretical impedance than speaker cable, and the graphs above show exactly that.

It is good to see impedance matching to a load, but other variables are in play, and impedance matching isn’t meaningful or practical at these frequencies and impedances. Good designs usually address ALL parameters, however.

Interconnect and speaker cables, with VERY low audio range VP values show a much faster VP in the RF band. The values of 87% VP @ RF are NOT really correct for WHERE the cable is used, but “sounds” exciting.

What do we see at RF on an ICONOCLAST interconnect cable? We can calculate what we measured in the graphs above. We can use a grossly simplified equation to predict the VP based on capacitance measurements;

101670/(11.0 pF/ft * VP) = 105 ohms @ RF

Solving for VP we get a value of 88%, using the measured values of 1 KHz referenced capacitance. This VP factor will DROP considerably in the audio range to much LESS than that. Imaginary values (L and C) stay the same from 1KHz to RF frequencies so VP is changing;

101670/(11.0 pF/ft * VP)=~2000 ohms @ 100 Hz (audio)


VP = 4.3% @ 100 Hz. (101670 / 2156 Ohms * 11pF)

VP = 57% @ 20 KHz (101670/163 Ohms * 11pF)


VP = 2.17% @ 100 Hz 101670/ (278 Ohm * 45pF)

VP = 55% @ 20 KHz 101670/ (41 Ohm * 45pF)

If we take the VP reduction factor of a coaxial cable into the audio band @ 22 X lower, we see; 87% / 22 = 3.9% @ 100 Hz. Close to the same answer in our rough calculation.

The data shows a 13X to 20X or so DECREASE in cable speed as we drop in frequency. Signal arrival times are NOT staying in perfect symmetry relative to the input start point. The AMPLITUDE may be near the same, but the TIMING is certainly not. Arguments persist as to how long the cable needs to be to her the arrival time coherence.


The issue – All cables should terminate into their characteristic impedance (not really true at audio). At audio, the cable isn’t a fixed impedance, or even really an “impedance”. Interconnects see a resistive “infinite” load, but not speaker cables, which see a highly reactive low impedance load. 

Impedance is a REACTIVE vector value. This is a dead giveaway that we’ll have to deal with Dv/Dt stuff. All cables are a wire that is in series with an inductor and a capacitor to ground. All three R, L and C, keep getting bigger the longer the cable on a bulk value basis. The impedance is a VECTOR sum of the REAL part and the IMAGINARY part. The PHASE is created by the imaginary part of the impedance vector value. The impedance values aren’t the same for all frequencies (see the 1 KHz and 1000 KHz chart below) since VP keeps changing, and this is a component of the impedance value. Since the impedance is a vector sum magnitude ratio, it stays constant for each frequency point no matter how long the cable is. R, L and C increase proportionally.

      Reactive Change with Frequency

Most of us kind of know that we are supposed to match the impedance to the load for the best transfer of energy. We are actually only terminating the resistive component we call “impedance” to the load; a resistor in the case of interconnects, or a speaker load for low-impedance speaker cables.  There is a reactive component that is also at issue for good signal transfer. That reactive (usually capacitive) part of the Impedance vector magnitude diminishes the transfer of energy in time. Audio is not RF, so this matched resistor to resistor ideal isn’t exactly correct anymore, even for high impedance interconnects. The physics of the velocity of propagation make impedance matching impossible at audio as does the wavelength, which is far, far too long to react like a true “impedance” vector.

For transmission line effects to be a factor, the cable length also has to be at least 10X or more the quarter wave length of the frequency of interest. This relates to the fact that a voltage change has to happen BEFORE it gets to the end of the cable and audio speaker cables transit times are too fast, even @ 50% VP, for this to happen.

A cable can have impedance (real and imaginary values), but it is largely irrelevant to true load matching. There can be a signal reflection based on the CUT length of the cable relative to the speaker. This simple reflection can be absorbed with a ZOBEL network across the speaker terminals if it induces amplifier oscillations. But, low cap cables are benign to amplifiers, even with this simple length defined reflection.  The cable will sound the same with or without the network as the parallel circuit is not in the signal path. The tertiary effect of better amplifier stability is what improves the sound with too high capacitance cable.

At RF, a signal is “used” efficiently only when two like resistive loads see each other. RF cables are designed so that the cable impedance matches the restive termination load. Audio cables don’t work like this at such low frequencies since we can never transmission-line “impedance” match to a load with short passive cables. But, the “work” done across the load STILL has to be resistive. The imaginary components of a vector (Impedance is a vector sum of the real and imaginary components) store and release energy since they are composed of reactive variables; Capacitance and Inductance, both variables, are store and release variables of voltage and current respectively. Short cables still have reactance.

We can see what happens at RF. The graph below shows actual cable data of what is called Return Loss. The return loss, RL, represents the “reflected” signal that does not transfer to the load for an RF Ethernet cable. RL= the imaginary part that can’t do work till it is “real” or resistive. Notice that we see several RL values “dead nuts” on 100-ohms from a low of -55 dB to a high of ~ -22dB. WHY are the RL variables not all the same? The impedance shows 100-ohms for all those RL values. The impedance at every frequency has a different reactance due to a lot of things too complicated to explain today. Simply put, at the frequencies with the lowest imaginary component, more energy is transferred to the load. In our example, if the impedance is above or below 100 ohm, and more or less reactive, the RL is decidedly worse. This is the cause of the FAN shaped graph that we see below.

Audio cables aren’t used at RF, though, and suffer from simple reflections more than load matching ones. This isn’t bad thing, as the critical attributes at RF aren’t restricting what we need to do in the audio band for better signal quality. We don’t need to worry about minute wire diameter fluctuations that cause the above graphed RL reflections. Audio wavelengths are too long to see the diameter variation issues so designers can work with geometries that may not be ideal at RF, but are far more useful for coherence adjustments in the audio band. Those adjustments still have to be real, of course, and measured or calculated with accepted standards.

Audio speaker cable with AC signals is terminated into a load that is resistive and reactive. Alternating current reacts to the imaginary circuit cable values and regulates how fast, and when, we can get work out of the cable. Some early cables were so reactive that amplifiers would shut off using them. Even though our cable is not a true impedance we do have reactive elements.

Interconnects see an “infinite” ideal resistive load; 47K-ohm on up, and speaker cables see a very low, and varying, reactive input impedance (the impedance of all loudspeakers changes with frequency).

Speaker cables are CURRENT signal devices that are designed to transfer power to an electromechanical motor. And, a motor that constantly “changes its spots” at every frequency as does the cable. The “argument” between the speaker EMF and cable is complex. 

Interconnect cables are VOLTAGE signal devices terminating into a HIGH impedance resistor. We want to transfer the signal shape and amplitude to a load. To avoid distortion(s) we don’t want the cable or the load to mess with the transmit circuit, but they do.

Audio cables are way too short to be transmission lines, needing at least 10X the wavelength inside the dielectric to be a true transmission line. Even 20 KHz is way too long a wave length to match that definition. We DO have simple reflections off the LOAD (speaker itself) that cable can’t manage as the load varies with frequency. This is very different than RF where I can make a cable nearly look like the load, minimizing reflections. I said “nearly” as all cables exhibit reactance, a TIME based storage of energy. Audio cables have significant measured time based propagation error due to VP and now we add-in a rise time error from reactance. The reactance of cable can be used to calculate “time constants”. At audio every frequency is associated with a different constant value. We’ll look at time constants later.

Zobel networks have been used to good effect to dampen the cable to speaker load variation, but they are estimations of where the two are most aggressively reflective. A Zobel network is a passive means to connect two differing but fixed characteristic impedance lines with a resistive value. Neither the cable nor the speaker are linear loads making it an approximation as to where to tune the Zobel network.

For more on Zobel networks and speakers go to;

Zobel networks and loudspeaker drivers

Compared to our “typical” Belden cable (blue trace), ICONOCLAST is flatter (orange trace) in velocity change as we go lower in the theoretical impedance. This is more the result of a higher, but still low, capacitance between the two designs. Lower inductance was preferred over capacitance.

The table data below is REAL and represent what even really good cables do through the audio band. The physics of the propagation delay match the measurements.


The interconnect tables follow and yes, they too show time based changes.


            The issue – What do the reactive variables do with respect to frequency?

Capacitance and inductance are essentially FLAT with frequency. Yep, capacitance and inductance are, interestingly, the same from near DC to the “sky is near the limit” frequencies.  Capacitance is set by the dielectric, assuming it is a linear dielectric material, and some aren’t (PVC). Measurements show that stable dielectrics offer frequency linear capacitance. Inductance is set by the distance between the wires and the loop area; it isn’t changed by the dielectric at all. These two values are always steady Eddies, but their time based effects on current and voltage change with frequency.

       Here is ICONOCLAST speaker cable that shows L and C data,

and it is FLAT with frequency using Teflon® as the dielectric.

The choice of what plastic to use sets the dielectric constant. You want stability with respect to frequency. Teflon® has the lowest dielectric of any SOLID plastic and thus the lowest capacitance with the thinnest walls of any material and, it is durable. It costs a LOT to buy and process, too.  Cost isn’t why we use it, performance is.

Plastics aren’t magic for capacitance, that is just the way it is. You want to pick the lowest dielectric constant value not just for low capacitance, but to help offset the change in the dielectric constant with respect to frequency. PVC dielectrics are far worse in linearity with respect to frequency, and the slope is not the same everywhere.  The chart and graph below assumes a set wall thickness and changes to the dielectric material alone. We can alter the WALL thickness based on the dielectric constant to get a given capacitance between two wires. Double the dielectric constant means doubling the wall for the same capacitance. Use the cheap stuff then? Sure, but more wall thickness increases loop area (space between the wires) which increases inductance! Oops, we’re not going to get zero cable reactance that way! A wire in a vacuum inside a braid ground would be the smallest size with lowest capacitance you can realistically see. This design would also have the lowest inductance since the loop area would be at a minimum with the vacuum acting as a low dielectric material.

We can calculate the effects of the dielectric and capacitance using a shorthand RF formula 101670 / C *V. We fixed the reactive impedance to a set value, so for a fixed wall of insulation, the capacitance rises as the dielectric constant is higher.  Since we know that the capacitance value is flat with frequency, this applies to the audio band as well. Better dielectrics for a given wall mean lower capacitance. This has nothing to do with Inductance, which is related to the magnetic field lines. Inductance is related to the distance between conductive surfaces, the less the better and field cancellation…if any.

Impedance = 100 ohms

Velocity = 1 / SQRT (E)

Capacitance = 101670 / (impedance * VP)


The issue – all cables store and release energy (current or voltage) reactively to the frequency being electromagnetically moved through the wire, adding time based distortion.

Look at the Impedance / Phase trace shown above on Part 3, Velocity of Propagation Issues. Notice that the PHASE on BOTH cables changes. PHASE include reactive components that TIME shift the signal’s ability of a signal to become resistive (If the phase trace hits “0” the circuit is resistive and has no reactive component).

A capacitor looks like an OPEN to DC or to very low frequency AC voltage changes. The cable is very “reactive” to voltage changes at lower frequencies. As you go up in frequency the cable’s bulk capacitance looks more and more like a SHORT circuit. The cable becomes more “resistive” looking with less reactance to voltage change. The trace explains why we can use 75-ohms and 100-ohm loads for RF cables, they look “mostly” resistive at RF.

Using a high impedance probe to measure the cable’s reactance produces the traces that you see. There is little current flow into the cable and, this is essentially how interconnects are used. The terminating load is VERY high impedance limiting current flow. When you put a voltage across a capacitor (our cable) it sends a momentary inrush of current to try to fill the capacitor. Output devices loading the circuit are ideally super low impedance to allow for this “inrush current”. Cables with lower capacitance mitigate the inrush current issue. Current LEADS voltage in a capacitor so there is a TIME shift caused by the cable.

Speaker cables are differing in that we don’t measure them like they are used; terminated into what is essentially a short circuit, the speaker. The large current flow in speaker cables responds to reductions in INDUCTANCE. Inductors resist current flow changes and that’s what speaker cables are trying to “move”. Voltage leads current in an inductive circuit and again, we see a TIME shift caused by cable but the opposite reactive variable, inductance verses capacitance, than the interconnect cable. 

Also consider that in speaker cables, the most reactive region is exactly where speaker’s impedance is also the most reactive, too. We want is a cable that is purely resistive but that’s impossible since a cable is a vector of capacitance and inductance.

Can we look at this another way? Yes, we can. If we examine the capacitive reactance equations below, and stick in the values at DC (F=0) and infinity frequency (remove F) and see what the results are we get the same answer; reactance is high at low frequencies and lower as you go up in frequency.

Xc= ½ * pi * F * C

XL = 2 * pi * F * L

The inductive reactance is the opposite, it looks much smaller at DC (F=0) than at higher frequencies (F= infinity). An inductor is a SHORT at low frequencies and an OPEN at higher frequencies. Fortunately speaker cables are relatively lower frequency making things less severe than at RF.

Cables, and all circuits, have capacitive and inductive reactance. Capacitive reactance resists voltage change and inductive reactance resists current change. They are both frequency dependent.

The TIME it takes to CHANGE the signal applied against a reactive load is measured in TIME CONSTANTS. It takes about 5 to 6 time constants to reach steady state amplitude. Our signal is also distorted the longer it takes to reach steady state amplitude so it may get nearly as big (we’ll pretend attenuation isn’t an issue), but it isn’t the same SHAPE. Don’t forget, every frequency is associated with a different time constant, and the decay or removal of the signal is the inverse. It takes TIME for the signal to bleed away to zero and this alters the decay signal.

As frequency changes, so do the reactive variables the determine a cable’s reactive performance.

At the very high end of the graph below, we see simply SQRT (L/C). At the low end the simple reactance (denominator) enter in.


            The issue – Current magnitude normalization at audio frequencies. Is this real?

There are several ways to calculate skin depth, and they all will yield the same answer. Impedance / RL can be derived from several inter-related factors and so can skin depth. It is real, and it can be managed to control phase distortion.

We all know about skin effect, but WHAT exactly is it doing at audio frequencies and is it real? Yes, skin effect is real at audio and all industry accepted calculations show that it is. The definition of skin depth is the point inside a wire where the current decreases to 37% the surface current magnitude. Skin depth is always the same depth of penetration no matter the wire size. Skin depth will vary based on the material’s electromagnetic properties and the frequency of the signal. For audio we calculate @ 20 KHz.

µ = permeability (4π* 10-7 H/m), note: H = henries = Ω*s

π = pi

δs = skin depth (m)

ρ = resistivity (Ω*m)

ω = radian frequency = 2π*f (Hz)

σ = conductivity (mho/m),  note: mho [Electrical ‘mho’ symbol – RF Cafe] = Siemen [S]

At low frequencies it simplifies to;

δs  = SQRT (2ρ/ ωu)

Looking at COPPER, we would calculate 461um (0.0181” depth).

If the skin depth at a given frequency is 10-mil on a 100-mil wire the 37% current point is well near the wire’s surface, it’s just 10-mil away in 50-mil radius.  If we halve the wire size, the current magnitude is larger through more and more of the wire. Each time we decrease the wire size, the larger the current magnitude becomes across the wire relative to surface current. In our 18-mil skin depth wire example above, the current in the “center” of a 36-mil wire will see 37% the magnitude of the surface current. Making wire smaller will INCREASE the current magnitude in the wire’s center to be closer and closer to the surface current in the wire at higher frequencies.

AC resistance involves FREQUENCY which is a TIME based variable.

RAC= (RDC) (k) SQRT (Freq)

K is a wire gauge factor that involves skin depth.

Freq is in MHz.

The internal wire impedance (AC resistance) is driven by the INTERNAL magnetic field’s relationship to inductance. Inductors RESIST instantaneous current flow and have higher “resistance” as AC frequency goes up. Current flows in the least resistive part of the wire as frequency goes up, so it reaches the surface where the self-wire inductance is nearest to zero.

Once we flatten the velocity change as best we can with a good dielectric design, we need to ALSO time align the effects of the dielectric at ALL frequencies using SMALL wires. Small wire improves arrival times as it forces the effects of the composite dielectric speed to be more uniform, as best we can, at all frequencies. This counters the skin effect problem that moves the current density magnitude to the surface of the wire as frequency goes up. Smaller wire increases the current magnitude (arrow length) in the wire center region to make it more efficient at time alignment.

One BIG wire
More SMALL wires

Even if we have the SAME current magnitude throughout the wire at all frequencies (impossible unless our wire is one atom in size) the velocity of propagation of the electromagnetic wave energy is STILL different at every frequency! But the ears say if we MANAGE the problems, our cables can sound much better. I took the time to measure all of this and flattened the impedance trace as much as I could. The VP changes less with frequency the flatter the impedance curve. Capacitance stays the same at all frequencies, so this VP is therefore changing less the more consistent.  Smaller wires are more consistent dielectrically at all frequencies.

Bigger wires will cause even more signal speed change relative to frequency because each electron’s is far smaller inside the wire. Each magnetic field contribution changes velocity the closer or farther that electron is away from the dielectric material. When a current is applied (electrons start moving) an inner wire located high frequency current mode travels slower than the same frequency signal on the outer wire surface and all these current modes are superimposed one on top of the other. This is called group DELAY.

Not all signals at the same frequency arrive at the same time, it depends on WHERE they traveled (MODE path) through the wire and  what the velocity of propagation is from the geometric perspective. The lower in frequency you go the less you can change the group delay since the current density through the wire is more and more consistent.

The overall magnetic field is a summation and superposition of ALL the moving electrons, the whole “group”. This is also why air is often used in interconnecting cables to mitigate the dielectric’s impact on the signal, and why you see more small wires in speaker cables. Electromagnetic field uniformity in the dielectric is important. The overall audible improvements are more debated. But, there is science involved in the optimization process.


The issue – dielectrics can impact weak electromagnetic signals disproportionately. Electromagnetic fields are squared law fields, and are most influenced by dielectrics nearest the wire. Weaker electromagnetic fields are most susceptible to dielectric distortions and the group velocity is mostly set by the strongest signals dielectric medium.

Using too many small wires splits up the current and starts to allow the dielectric to influence the sound more and more, negating the “advantage” of dielectric uniformity. The electromagnetic field is strongest nearest the wire, decreasing with the square of the distance moving out away from the wire. The electromagnetic signal moves from being “in” the dielectrics to being around it. The signal propagation speed is an average of ALL the dielectrics, with the material the stronger fields reside in have the most influence on the average of the “group”.

Four-fifths or more of the current magnitude at audio is below 3 KHz. Some call this the spectral power density, or roughly where the most energy is being placed. The electromagnetic energy does not STOP in the plastic or air. It emanates out in an inverse LOG power decay THROUGH all the materials it encounters along the way. The predominant material VP effect occurs CLOSEST to the wire. The smaller the signal (interconnect cables) the bigger the effect of the immediate dielectric nearest the wire.

Weaker signals will be impacted by the dielectric’s effects more than stronger ones, as they decay to far weaker signals moving away from the wire. The speed is more and more determined by the dielectric near the wire as we go up in frequency. Interconnects see little of the plastic out away from the bare wires as the field decays so quickly, but, the smaller the electromagnetic signal, the MORE it is influenced by the material nearest the wire. That superposition of materials SLOWS the signal (air to plastic) or speeds it up (plastic to air) relative to just the initial material’s properties.

We can see this in actual practice as the “group” velocity of all the materials on Ethernet cable shoes a value SLIGHTLY higher than the dielectric (66%) itself, and measures 71%. The signal is in the “air”, a good dielectric” and this influences the overall signal speed.

ICONOCLAST interconnect design switches this around, and puts the AIR nearest the wire, where the signal strength is highest. This negates the outer plastic dielectric’s contribution to the group velocity, so we see a higher 87% value at RF. This translates to lower capacitance number where we use the cable in audio applications.

The VP speed variation caused by the “composite” velocity is complicated by the fact that the LOWER in frequency you measure, Mother Nature’s devious plan slows everything and this time shifts the audio band.

We can’t change the fully diffusion coupled (same magnitude current through the wire) low frequencies, so we try to time align the faster upper frequencies. At RF the upper frequencies are “on” the wire surface so the dielectrics affect them nearly 100%. At RF this is fine because it is near the same VP at all RF frequencies. At audio, we want to move most of the high frequencies AWAY from the dielectric so the speed is closest to the lower frequencies.  We already know that the VP is faster the higher in frequency we go so this messes up the signal arrival times.  The only good way to slow the upper frequency magnetic field is to make the wire smaller so less energy is JUST at the wire surface nearest the dielectric. More current is “in” the wire versus “on” the wire based on skin depth.

Can we overdo field current normalization? What if we could make a wire one atom wide? Now, the impact of the DIELECTRIC is as big as it will ever be and with a really, really small current in each wire. The total current will be the sum of all the wires we want to use in parallel. The more wire you use, the smaller the current in each wire. Current is the number of electrons past a point with respect to time. Well, we have ONE tiny electron moving in each “wire” and THAT is as small a current as you can have! Model a weak signal, and the electromagnetic wave is so weak it never really leaves the dielectric, whatever material the dielectric is. The dielectric better be really decent as it is hugely involved in capacitive rise time (calculated capacitive reactance rise times constants) signal arrival time (velocity of propagation).

At very high frequencies, and if the wire is infinitely big, we see ONLY the dielectric as the current is at the surface (skin effect). Likewise if the wire is infinitely small we AGAIN see JUST the dielectric (no skin effect can happen). Between the extremes of wire size, somewhere, we can alter the arrival time of the upper frequencies with wire diameter and dielectric choices.

Interconnects are much easier, but not real easy, as they terminate into a high resistance, nearly open looking circuit. The reflections off a CONSISTENT resistive load of 47K-120Kohm aren’t as bad as the mismatch speaker cables experience as BOTH the cable AND the load are in constant flux. Worse, the speakers change by design! The seemingly high measured impedance slope of RCA or XLR interconnects in the initial graphs aren’t as bad as they seem. Not only are the “impedances” not real at audio but you have far bigger issues with the non-linearity of cables loading the output devices in your preamplifier. Trying to match ideal infinite input impedance on RCA or XLR cable would mean tiny capacitance values. We go as LOW in capacitance as we can to allow the output devices to see an easy load. This is way we shoot for keeping capacitance reasonably low.

Massive signals in the speaker cable are less impacted by the “composite” dielectric speed. The electromagnetic field will travel at an “average” of all the stuff it is moving through, so the better the “average” material is that the electromagnetic field is in, the FASTER the signal travels, and the less TIME the signals have to become separated as they travel down the cable. This is the time and distance story problem.

PC’s stopped using FLAT cables because the signal arrival TIME differential got to be too high. They went to SERIAL digital designs, and re-clock the data from memory. This at first seems counterintuitive, adding the re-clocking circuit, but unless the TIME can be managed, you’re screwed. Faster is better but I’d take SLOWER and the SAME in an instant! This is the “keep cable shorter” thing, but to be LONGER we have to be FASTER, too, if time errors are to be kept low. Mother Nature says we get a raw deal in the audio band verses RF.

In speaker cable, the stronger low frequency electromagnetic waves emanate into the air through the plastic dielectric more than the higher frequency signals so they are theoretically aided by the air around the wire (superimposed dielectric value) more than the weaker high frequencies that see more of the slower plastic dielectric. But the VP erosion as we drop in frequency eats-up that advantage in the low-end. It’s there, but small. The problem is that the low frequencies still drop in speed way more than the air’s addition to the overall speed. Seeing more air as we go lower in frequency speeds the signal up relative to the faster high frequencies and offsets some of the problem…but it never aligns it away to zero. The VP still marches slower and slower as we go lower and lower in infrequency. Arrival times are more important than SPEED down the wire.

The highest frequency carried in the speaker cable is most fragile, but compared to interconnect cable, it is relatively robust.  The high impedance interconnect cables are yet another problem. ALL the signals are VERY, VERY low current electromagnetic field energy states. Here, I need the BEST material possible to time align the energy field “whipping” (slowly whipping) down the wire; air. The VP is the inverse of the dielectric constant so we want a fast dielectric and the lowest associated capacitance it can also provide. This is why I HAVE to use AIR core designs to properly time align the energy AND use SMALL wires to better distribute the dielectric’s effects at ALL frequencies nearest to the same composite velocity. The third leg is to decrease output device capacitive loading. Air helps mitigates velocity variation across the frequency band that is the bane of audio signal transmission. It incidentally also pushes UP the “impedance” to better match the load, the opposite of a speaker cable. I’d be wary of that improvement as we need to be aware that audio isn’t a transmission line.

It seems counterintuitive to use air, as it speeds up the higher frequencies relative to the lower frequencies (makes the difference worse) but the capacitive reactance influences rise time error if you let it get too high. The propagation time and the rise time need to be balanced, somehow. There is no perfect solution.

We call, it “sound quality” when we use the cable, but it really is the arrival time alignment of all the signals. The human brain hears superimposed time alignment and amplitude preservation first, everything else a distant second. The argument is: does this make a difference?


The issue – Changes in electromagnetic properties between interconnect cables types can alter the ideal “tone” that was intended.

An often ignored issue is, what do you do with a really good sounding RCA cable? Why not make a really good sounding XLR that’s the same reactive measured design? Most RCA to XLR cables never match. ICONOCLAST is no accident. I purposefully designed the RCA and XLR to be the exact same reactive match and thus the same “quality” of sound through the channel. The above impedance chart that we saw earlier shows both the RCA and XLR. Look closely, they are electromagnetic buddies.

Does that make a difference? If you have a very good RCA design, it sure can’t hurt to start there on the XLR!


The issue – how to make complex cable’s cross section look like one simple wire electrically, and every wire sound the same?

Matching multiple wires into a complex structure isn’t easy to do well. The ideal cable is one wire that is exactly the same as the opposite polarity wire. To meet other objectives, we usually have several wires.

More small wires will make a nice big capacitor (wires with a dielectric between them) and trash reactive signal conversion to resistance products. Inductance will inversely follow capacitance, messing up the current delivery, and to get BOTH intrinsically LOW, you can’t go “whole hog” on the opposite variable. The two variables are tied together inversely. Rats! A suitable compromise must be reached? Yes, audio is a compromise, as we are seeing. BOTH L and C need to be low in value and good design manages this. Trade-offs for better sound can, and should, be logically explainable.

The less understood variable is Inductance. This variable is a big contributor to more wires. We all think, “capacitance” for audio. Realize that if we had NO inductance, we could separate the wires as much as we wanted and eventually have no capacitance (outer space actually has capacitance, so that’s impossible, too). The reduction of the magnetic fields by proper cable geometry reduces the inductance, allowing a larger wire center-to-center distance for low capacitance.  The LOWER the electromagnetic field, the LARGER the loop area can be (lower capacitance) for a given inductance and vice versa. Too many cables ignore getting the electromagnetic field as low as possible. The higher the current (speaker cables) the more field energy you need to eliminate.  Wires with low electromagnetic fields and small loop area have the lowest inductance.  ICONOCLAST’s speaker cable design balances out the wires’ proximity to one another so as to not “rob Peter to pay Paul”. The unique weave pattern increases the average wire C-C (Center to Center) distance creating a wire pattern that CANCELS the electromagnetic field while increasing the average spacing for low capacitance. Cancelling the field energy allows me to also lower inductance which would be impossible to do with JUST wire spacing for capacitance alone. No magic need apply.

EVERY wire in a cable has to be the same wire if you use superposition of the electromagnetic fields traveling down each wire. This is why symmetrical cable designs are used to efficiently remove reactive time alignment issues. I measured the reactive time based issues on other designs and they all came up short. Capacitance and Inductance have to be the same on EVERY wire to as tight a manufacturing standard as is possible. Multiple, and differing wire sizes are too complex to align things nearly as well. The signal’s SPEED has to be best matched at all frequencies and not just the physical wire length. The wire’s “signal length” is the problem. Use too many non-symmetrical, differing sized wires and this is all but near impossible to do with all the variables involved. I call this type of mixed wire cable, “cable in a cable”. The effect is a kindergarten lunchroom in the dark; a mess.

In passive cable, you can’t force the highs to go in the small wire and the lows in a bigger wire, and adjust the wire lengths to offset the VP changes. The ENTIRE spectrum goes into EVERY wire, so now we compound the time based issues. Only active electronics can separate the spectrum, and that’s a problem too.

Does ICONOCLAST remove the “cables in a cable” problem? Only one way to find out and that is to MEASURE them. The data is showing each polarity with 12 two wire BONDED pairs, 24 wires in each polarity, and 48 total wires in each cable;

175.3815 pF (X) +175.3815 pF = 180.1803 pF

175.3815 pF (X) + 171.4507 pF = 171.4507 pF

X = +2.74 % and – 2.29% variation between wires, or, they are ~ 97.5% the same.

I would say yes, I got it right.

For ICONOCLAST speaker cable I set my design goal at no more than 50 pF on capacitance and 0.1 uH/foot an inductance (45 pF and 0.08 uH typical). On the interconnect I set the goal at 12.5 pF and 0.16 uH (12.0 pF and 0.15 uH typical).  This is WITH connectivity and tested to prove it.

The complex electromagnetic designs of the RCA, XLR and speaker cables allow ICONOCLAST to exist. The RCA is the most pure electromagnetic equation that I have to work with and defines the interconnect cable problem. How do we reach the greatness that a PROPERLY designed RCA does in the XLR design (matched impedance / phase)? How can I convert the small signal world of the RCA and XLR into the large current world in the speaker cable (low inductance with still low capacitance)?


            The issue – is it mostly LOG linear so we can’t hear it?

If it is true that we can’t hear LINEAR attenuation (measured Rs values say there is non-linearity) or TIME based issues in audio cables, WHAT are we hearing with optimized designs, i.e. those that try to get L and C to near ZERO as we can and with low time based issues? The design goal difference in ICONOCLAST is TIME based and I’m not so sure that the inaudibility of difference values of 5-10 micro seconds is correct. Linear attenuation, I agree, is MUCH harder for the ear to pick out in typical cable lengths. I said LINEAR LOG type decay.

Rest assured, if there is snake oil in these products it sure looks like physics to me. All the above data is measured and real. The question remains, WHY do the cables SOUND so much better if TIME based issues aren’t audible? WHAT are we hearing, then? The reactive TIME altering L and C along with the VP change with respect to frequency seem to be the difference in cables, and audibly so. Linear attenuation can’t account for the differences. Series resistance says that that factor isn’t as linear as we’d like, either. There is a measurable difference in cables resistance across the audio band.

Is attenuation linear? I measured the Rs (series resistance), with respect to frequency, of ICONOCLAST and saw a significant CHANGE in attenuation with high quality R, L and C. Look at standard 1313A speaker, 10 AWG Zip cord style cable (red trace). ICONOCLAST flattens resistive non-linearity artifact, and the interconnects are both flat to 20 KHz human hearing test point.  Still, look at the UNITS; it isn’t a wall of lost energy above 20 KHz.



12.0 Low Pass Filter Effect

The issue – Cable is a low pass filter, and rolls off the frequency at the frequency of the filter’s cut-off; Fc.  How does this change what we hear?

I saved this one for dead last since it was even overlooked on my categorization of issues with audio cables. You’ll see why in a moment.

Here is the basic circuit. There is actually a SMALL inductor in series with the resistor but notice that it doesn’t appear in the equation that defines how the filter will behave, and is omitted. There are circuits that involve larger inductors, and unless we have a resistor to ground, they won’t apply to “cable” filters. Well, decent cable anyway.

The capacitor is a reactive device, like I’ve mentioned before, so its properties change with frequency as does an inductor. A capacitor eventually looks like a short to ground (capacitive reactance value keeps changing) at higher frequencies so the signal energy takes the path of least resistance through the capacitor to ground. R is in Ohms when capacitance is in picofarads (pF).

The good thing about almost ALL audio cables is that the roll-off properties of the filter are WAY above the audio band. Yes, a first order filter will change the PHASE at the -3dB attenuation point by 45 degrees, and time based distortions are more audible than the roll-off attenuation. First order filter attenuation nor phase changes are going to be an issue, theoretically.

Typical ICONOCLAST™ R, L and C Variables

Capacitance12.5 pF/ft12.5 pF/ft45 pF/ft
Inductance.15 μH/ft.15 μH/ft 0.08 μH/ft
Resistance32Ω/Mft 14Ω/Mft 1.15Ω/Mft

The RCA shield “goes away” as it is such a low resistance in series with the center wire, leaving essentially the center wire DCR.

The XLR uses TWO 25 AWG wires in parallel for each polarity, so the resistance is HALF the two wires, or about the same as a ~22 AWG wire.  

 Calculating Fc we arrive at;

15.5 GHz for the 5 foot RCA.

36.4 GHz for the 5 foot XLR.

15 GHz for the 10 foot Speaker Cable.

The real problem with cable is that it can load down the output op-amps with too high capacitance and change the frequency response and possibly phase response. Some really high capacitance or high inductance speaker cables can bug the heck out of power amplifier output stages, too. But these problems aren’t filter problems, but bulk capacitive or inductive loading problems on the output circuits.

All circuits “push back” below their operating region into the pass band but a rule of thumb is to keep the fc pass band 10X or more above the circuit’s operating frequency. We surely are meeting that requirement with any decent cable, even zip cord.


Many outside this sub-discipline of engineering will STILL insist that electromagnetic field time management and time alignment are not important, and that only the bulk R, L and C matter. The ear is a time domain instrument and readily time aligns the signal to the natural world we live in. EVERY effort was made to pay attention to TIME domain issues in audio cables and attenuation non-linear artifact. There are a myriad of ways to lose track of TIME, and an audio cable is not a good place to make mistakes.

Consider all the measured and factual information above on cable design and then ask yourself why cables sound different. Why wouldn’t they sound different given how complex  it all is? True, poorly made cables all fall into a bunch of warm and soft sounding products. Elevate the engineering and they indeed measure different. The above is 100% true for ALL cables, if I may add. If I mischaracterized a topic then, of course, only my cables are affected! All the cable designs in the ICONOCLAST line are under US patents.

I hope my cables bring years of enjoyment to you, and NEVER a feeling of complacency in what was provided to enhance your hobby’s (mine too!) pleasure. The search is constant to try to align TIME based issues to arrive at the best sound possible. The bad layers of the onion can’t be removed, but the order and thicknesses can be altered. Signal coherence is both arrival time and amplitude time dependent. Passive cable won’t allow perfection, just a lot of hard work to manage the ill effects that Mother Nature threw our way.

“Sound Design Creates Sound Performance”, and this means driving down all measurable variables to the lowest possible balance we can achieve. Does this make better sounding cable?


Galen Gareis
Principal Product Engineer
ICONOCLAST Design Engineer

Shields and Noise Cable Dynamics

We tend to believe that SHIELDED cables are superior to UNSHIELDED cables but the opposite is true from a signal transmission evaluation. Why we feel SHIELDED is better is because we overestimate NOISE ingress (outside the cable into the cable) environmental issues.

Cable electrical is determined by the primarily REACTIVE variables that change signal shape arrival times. SHIELDING is to be considered a necessity if, and only if, the ingress noise is more damaging than the time based errors and physical size shielding imposes on cables. Why even have shields if it doesn’t HELP improve the signal integrity from one end of the cable to the other?

Capacitance is derived by the relationship of the shield to the signal conductors in cable. The shield is usually at GROUND potential to be a low impedance path for noise, so far so good. The bad news is that the CLOSER a shield is to the signal wires, the more the cable varies per unit length in measured electrical values of capacitance. It isn’t the same cable all along its length from shield geometry variation, and the variation is much more aggressive the closer the shield is to the signal wires. Capacitance, and thus also inductance, change with smaller physical changes in the cable.

If you want to keep cable size SMALL, a shield means much higher CAPACITANCE. And, a smaller size WITH that higher capacitance means a larger per unit length variation in measured electrical.  Even with AIR as a dielectric, we will see much higher capacitance, and have a harder time controlling it with shields, so we better need one for the function of the cable, and where it is used.

The following calculated table shows that the DIELECTRIC in-between the shield and the signal wire can REDUCE capacitance, but only to a point. It cannot remove the shield to conductor physical variation, which is built into the DESIGN, good or bad.

How bad is the actual variation between the shield and signal wire? This exact question was discussed when ultra high-speed communications cables were being developed. Do we control the center-to-center spacing in a BONDED PAIR over all else, or do we control the shield spacing and geometry AROUND that bonded pair? BOTH will influence the final impedance, and its variation. Which is really the bigger problem? Can we make better cables managing what really makes the biggest difference, and reserve the less aggressive physical attribute for higher performance requirements? This can make the AVERAGE level of performance much higher at a much lower cost than blindly trying to manage every variable all the time without a firm reference to the cable’s final electrical values and variations.

Here is that exact analysis;

To demonstrate the effectiveness of conductor center to center (C-C) in an ISTP cable, the example below shows a change of C-C from 0.055” to 0.072”, holding a constant 0.061” insulation diameter. This simulates a conductor with poor concentricity within a well-controlled and constant insulation diameter. The impedance is nominally 102 ohms with a 0.061” C-C spacing and changes ever so slightly as the conductors are spaced closer, or farther apart.  The shield inside dimension is a constant 0.122”. Under these circumstances, the impedance goes from just over 101 ohms to just over 97 ohms. A total impedance spread of about 4 ohms.

The significance of the calculations is the relative insensitivity of impedance value with changing C-C spacing compared to the variation in diameter of the shield, both of which affect impedance variation with frequency. The impedance versus shield spacing graph shows how severe the impedance change is with ISTP shield inside diameter (I.D.) changes. Just a 20 mil change (0.120”-0.140”) moves the impedance almost 14 ohms. Our specifications allow only a 15-ohm swing. 

The control of the effective shield diameter is three and one-half times more sensitive than the C-C spacing of the conductors in ISTP cables. Or, shield tape control is much more important than insulation centering or backtwisting to compensate for off-center conductors. Also notice that the closer the conductors move towards the shield in the IMPEDANCE VS CONDUCTOR SPACING chart, the more Zo changes. When the conductors are 0.055” to 0.065” C-C, the impedance varies by less than one ohm. In contrast, when the conductors are near the shield in the 0.065” to 0.072” C-C range, the impedance changes 4 ohms. Unless your C-C is well out of spec (we have a 0.01” variation with little change in impedance in this example) good shield dimensions are much more important.

In contrast to ISTP cables, the UTP cable example shows how profound the impedance impact is when the C-C changes just 11 mils compared to 17 mils in the ISTP example above.  Where the ISTP cable had about a 4-ohm swing, the UTP cable has a 60-ohm swing! In UTP cable, ground plane consistency is inherently stable because it’s the metallic area around the cable which, under normal circumstances, is perceived to be infinitely far away by the cable, too far to effect the electrical to any significant degree. So the crucial variable in UTP cable for consistent impedance is the strict control of C-C. This is why Belden’s patented bonded pair technology is so important in UTP cable designs.

Impedance is, after all, a function of the Inductance, capacitance and dielectric values. The impedance variation, and even at each frequency in the audio band, changes with the dielectric and the spacing.

A cable with NO SHIELD, sometimes called UTP, does have a reference ground “around” the cable, the environment. But, the capacitive and / or inductive coupling are so far away that changes in the “reference” are essentially zero.

Shields have to pull their weight in signal integrity improvements compared to cables used without a shield. If we have no external noise, SHIELDS ARE WORSE than no shields! The math of cable electrical stability firmly squares that up, per the data shown above.

This forces the consideration of NOISE. It even considers HOW noise is transferred into (ingress) a cable, and even if the cable itself is the source of NOISE for other external devices (EGRESS).

First, let’s be super straightforward about this from a 25,000-foot view. The closer a shield is, the capacitance value is high, and it varies the most around the average value. Knowing that the proximity a shield has around the signal wire can really upset the cable’s uniformity of electrical, and how uniform we can engineer them, would we not want to use designs that NATURALLY calculate an advantage to use with shield? Yes, we would.

To keep this easy, look at coaxial cables. This technology HAS TO HAVE a shield to work. A signal wire surrounded by a shield. The signal waveform travels along the wire surface, and under the shield surface and inside the dielectric as a TEM (Transverse Electromagnetic Wave) wave. The more perfectly round the inner surface of the shield and the outer surface of the signal wire, the lower the capacitive and inductive variation and thus a lower impedance variation.

For signal transmission, we use 75-ohm cable (77-ohm is the ideal) and for power 50-ohm Cables  (30-ohm is the ideal). Approximately 53.5-ohm military RG cables came about because it is the mean between 33 and 77. If we freeze the materials we use to make the cable (same plastics and metals) we will see that a 75-ohm cable has a larger dielectric layer (lower capacitance) than the 50-ohm cable.

This is nice, since the farther away the shield is, the less a given VARIATION of the shield changes the electrical stability. Reactive variation impacts small voltage signals far more than larger 50-ohm power cable applications with much more robust signal levels.

In a 50-ohm power type cable, we have a shield that is far closer to the signal wire. This seems like a problem and it is, but the SIZE of the signal is vastly larger than the NOISE. We can overcome the noise with a larger signal, and even the return loss caused by more variable impedance can also be mitigated with the size of the signal on power type coaxial cable.

This is simply the signal to the noise reference working in our advantage in each design.

  • Voltage signal cables, dB or dBm, need shields farther away (higher impedance) and it so happens this is the case with 75-ohm cables, reducing capacitive coupling of noise.
  • Power signal cables, often in WATTS, need closer shields for energy transfer (lower impedance) but this allows more capacitive noise coupling. 50-ohm cables use more robust signals to overcome the noise. This is like a low impedance speaker cable’s signal WAY over the terrestrial noise floor.

There is NO EXCEPTION, lower impedance cables are much more subjective to NOISE than higher impedance cables with the same noise ingress. We must fit the signal levels to the impedance for ideal overall performance. 75-ohm cables are far better for low-level signals as they capacitively couple less noise, as the DISTANCE to the shield is larger.

To put the signal in perspective to NOISE, look at the table below.

Digital data cable go 100 meters / 328 feet with over 23 dB of attenuation at 100 MHz and with ZERO errors due to external noise, with UTP designs. Audio cables go mere feet, and yes we seem to want to be the underdogs of signal integrity but we aren’t, and that’s a really good thing, too.

The integrity that even a MC phono cartridge’s 0.35mV signal represents to the noise is in our favor.  The robust signal even covers up POORLY made SHIELDED cables. Do the shield really right, and it can help some RF, but usually in a good unbalanced RCA system a RF bleed capacitor routes RF to ground through the cap somewhere in the ground.

Coaxial cables need shields to work, and they need shields to be super low DCR to prevent ground loop differential currents between devices. The GROUND is shared in coaxial cables at uneven ground reference points. RCA grounds have resistive differences. This can cause signal bleed between channels. A BIG part of an audio coaxial cable shield is to mitigate ground potential differences, and not to “shield” ingress.

A balanced XLR uses a SHIELD, yes, but it is NOT a part of the signal path, and each right and left signal doesn’t share the virtual ground between the differential voltage signals. Each amp has its isolated virtual SIGNAL ONLY ground reference. There can be no inductive or capacitive coupling of right and left channels. Unhook the GROUND on a XLR and it will work, with MAYBE a slightly higher SN ratio. The outer shield simply knocks down the noise ingress at RF, if any is there, so UNBALANCE in the pairs mitigates to a lower residual value. One-percent unbalance of a small signal is better than one-percent of a larger signal.

This is the true advantage of XLR cables over RCA. Both have good RF noise immunity with the XLR having far superior signal channel isolation and…low frequency noise isolation.

Since an XLR FLOATS the virtual ground independent from any other signal, noise is the same on each leg, so it cancels. We see the “difference” of each leg as the signal, which doesn’t change potential. This includes magnetic and electric fields. Coaxial cables can’t shield magnetic fields since copper is “invisible” to 60 Hz magnetic interference.

ICONOCLAST™ uses SHIELDS, but the WAY we use shields insures geometric consistency to the signal wires. Care was taken to insure a good BALANCE within the XLR signal wires so even if a shield is broken, little performance impact will be measured;

Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture, all tests performed on a 20ft specimen.

Pr to Pr(star quad) – 10.4113 pF/ft

UnBalanced:  Pr 1 to Shld – 401.9868 pF/20ft

                       Pr 2 to Shld – 405.9738

Cap UnBal ((diff/max) * 100) – 0.98%

Requirement – 3% maximum

SHIELD TRANSFER IMPEDANCE – This is a measure of the cable’s shield impedance in milli-ohm/meter. The lower the transfer impedance at a specification frequency the better the shield at that frequency. It is frequency and design dependent. The current traveling in the shield times the transfer impedance produces a interference voltage product to ground in the shield, E=I*R where R is the transfer impedance.

SUMMARY – Shields have to be considered relative to noise and the resulting S/N ratio since the application of a shield is ALWAYS a negative variable to signal integrity that has to be over weighed by true noise mitigation. If noise is paramount over the signal, then shields are a necessary requirement. If shields are a part of the signal path, then the noise they generate has to be mitigated with shield DCR.

ICONOCLAST uses shields properly, and insures that the negative influences are geometrically stabilized, and measured for performance in both RCA (DCR) and XLR (unbalance percentage).