Shields and Grounding

BACKGROUND: There is always discussion on how to ground a shield. The answer lies in what the worst-case noise situation is. It isn’t always the same answer. Do you leave one end open or ground both ends? If you ground only one ends, which end?

We can look at how the interference behaves to answer these questions and they follow understood electrical properties within swept frequency regions.

BODY: The first step is to understand what a shield is doing, and how. In the simplest terms, a shield creates two separate “electrical” environments, one on each side of the shield. One side is measured as a RATIO of the field’s intensity relative to the other side, in dB.

The shields we are working with are ONLY effective with an EM wave that is predominantly “E” field in nature versus “B” field, or magnetic. Magnetic fields are not shielded or blocked with conductive shields, but need a shield material that blocks magnetic flux lines or a low permeability material…think “material a magnet will stick to.” Those kinds of material allow magnetic flux lines an easier path than through air. We can capture and re-route the flux lines in low permeability materials. For this discussion, we will look at mostly electric field shields, stuff that conducts electricity.

What is an ideal electrical shield? It is a shield that 100% blocks electrical energy at the surface of the shield, and that has infinitely low resistance. Shields don’t have infinitely low resistance and they block electrical energy at differing impedance based on the skin depth of the shield at a specific frequency.

We can measure the effectiveness of a shield across frequency with a transfer impedance plot. This is measured in milli-ohm/meter. It describes the resistance we can expect a shield to have, and thus the ratio of the fields energy in the shield based on the CURRENT the resistance causes to flow at that frequency…and it is not linear.

The graph below shows the frequency-dependent nature of a shield. A perfect shield would have NO RESISTANCE and both ends would be identical and thus seem like a SINGLE point of reference to a flow of current. Since we have zero resistance across the shield, we can’t have current flow caused by the shields. We CAN have current flow between the two points connected at the ends of the shield. In a “perfect” world the GROUND at both ends is the same potential and thus, an ideal shield has ZERO current flow, and is a measure of the POTENTIAL on one side of the shield relative to the other, in dB.

Since we don’t have a perfect world, Transfer Impedance describes what to expect at frequencies based on the shield’s impedance, and how that shield resistance creates a CURRENT flow and thus a voltage (shield resistance times the shield’s impedance = a voltage). When we have different resistances at each end of a shield we have current flow.

Shield Type5 MHz10MHz50MHz100MHz500MHz
Bonded Foil +60% braid2015112050
Quad Shield 60% +40% Braid
Tri-Shield+80% Braid10.60.10.22
Bonded Foil +95% Braid10.50.080.091

The chart below is what JUST the shield impedance looks like for a set length of cable at lower frequencies. We see the same non-linear behavior of shield and frequencies.

A SEED (Shield Effectiveness Evaluation Device per IEC 61196-1) test shows the dB relationship to a Lower Shield Impedance. Series number 5 with a 95% coverage 45-degree braid and Duofoil tape is clearly superior.

Okay, we can see a shield is not perfect, and not linear. So how does this say what to do with each end of a shield? We have to weigh the CHOICE of HOW the shield WORKS to decide our fate.

–     If you have ideal grounds and meet IEEE bus bar grounding (see the 568C.2 or later grounding specifications) limits it means BOTH ends can be grounded and the shield current will inductively couple less interference than the shield ATTENUATES through its material composition.

–     If we have severe ground differential, we can induce a strong current in the shield that CAN, if the shield’s resistance is higher, induce noise into the core that is WORSE than if we disconnect one end. We convert our shield to an antenna, not a shield!

–     An antenna does NOT create two separate environments between them with the ratio of one measured to the other. One end of the antenna is infinite impedance (the open end) with the other end at ground. The antenna “wire” is as close to zero impedance as possible in order to NOT attenuate the antenna’s signal going to ground. The signal won’t go to the open end, but seeks the lowest potential in the circuit.

We trade the noise caused by a poor ground resistance potential between shield ends for the induced noise in an antenna’s wire parallel to the signal wires that induces a voltage based on the antenna resistance. In an antenna type ground, it is best to ground the SEND end, as the SIGNAL on the internal wires is as LARGE as possible relative to the antenna signal, improving the signal to noise between the two.

An often-ignored aspect of shields is HOW to ground one at lower frequency versus RF. There is a big difference and again, it is based on the shield characteristics at each frequency.

The charts below are derived from TWO slightly differing MODELS of RF shield inductive reactance resistance. I have this paper for those interested. But, the data is the same message in that as frequencies increase, the shield reactance goes UP. This necessitates a FULL 360 ground at the shield termination point in RF circuits. This is why good RF connectors are fully capturing the shield all the way around the cable. On your RF digital cables, use 360 degree grounds for the best true shielding.

SUMMARY – Most systems will have proper GROUND differentials between them and thus have near ZERO shield current. The shield relative to the signal wires will ATTENUATE outside interference. When you have poor grounds, it may be beneficial to unhook the “receiver” end of the shield and hope that the induced antenna current voltage is less severe than the induced voltage caused by differential shield ground potentials. This should be a SECOND choice, not the first. A properly working shield, by design, has a KNOWN shield dB rating that can be trusted in a proper electrical circuit.

An antenna ground’s induced voltage onto the cable is not fully described and is dependent on the GROUND proximity point and shield’s distance from the signal wires. In severe situations, it may be the best choice to mitigate noise to the lowest possible reference value as it is pretty hard to REMOVE a shield already on a cable. Some, such as coaxial cables, can’t be mitigated and need to be properly designed to EXCEED the ground differential by several orders of magnitude so as not to aggravate any ground differential.

ICONOCLAST will use double ground interconnect shields and proper DCR RCA grounds. Power cables should also use grounds at BOTH ends if you have a proper GROUND plane resistance such that ZERO current flows and thus you have ZERO induced voltage from differential current. An antenna type ground CREATES a differential in each end of the “antenna” by design (one end is ideally infinity the other is ideally zero) and is thus a second choice if you have known ground issues.

One last note, those heavy 10 AWG power or more cables, may provide benefit as they induce less ground differential resistance than smaller power cords as the ground wires are larger. The circuit may not need the power delivery of a larger cable, but the lower ground resistance values may be of benefit on longer runs in marginal power grid situations. This will improve a shield’s current to nearer zero across frequencies. The dB isolation numbers values are for a proper shielded system with IEEE and TIA compliant shield differentials.

The Transfer Impedance numbers are between two-reference point probes on a shield, and DO NOT need the ground potential differences for characterizations.

Speaker Cable Design Brief

Speaker cables are a very different animal than high input impedance interconnecting cables. A speaker cable connects to an extremely inconsistent 2-32 ohm (or even lower and higher!) reactive load created by the speaker. RCA and XLR interconnect cables see a much more consistent and resistive high impedance load making their electrical measurements far easier to predict. The speaker cable also suffers from the audio band’s velocity of propagation non-linearities seen in the interconnect, but also has to figure out a way to be LOWER in impedance to better match the speaker load, while the velocity of propagation is going DOWN, and this naturally increases the cable’s impedance. How is all this managed as best we can?  This paper is a walk through on how ICONOCLAST™ speaker cable addresses some of these issues.



  1. Conductors.
    1. Copper Size.
  2. Dielectric material(s).
  3. Dielectric geometry.
  4. Shield material and design considerations.
  5. Jacket design and material considerations.
  1. Conductors

For speaker cables, the first issue that has to be decided is how much CMA (Circular Wire Area) you need based on the application. This isn’t always an exact science as the cable length and speaker type will change your calculated answer. The speaker cable becomes part of the cross-over network in the speaker. The amplifier sees BOTH components as one load. 

Since the cable is seen as part of the speaker, it is easy to understand that the “reactive” relationship is between the speaker plus speaker cable and the amplifier.  Speakers vary by design so the overall speaker component back EMF portion of this load into amplifiers varies. Amplifiers of differing design react to the back EMF and the overall performance can be hard to predict. The goal is to “remove” the cable as best we can between the amplifier and speaker. Cables should not be tone controls, but that’s the goal of EVERY component!

The analysis below looks at the calculations that have been made to settle on the total CMA area for benign reactions to the frequency response of a typical set of loudspeaker loads. And yes, these are NOT real time resistive loads but as always, an approximation.

The general rule of thumb is that you want the total speaker cable resistance to be less than 5% of the speaker impedance PLUS the cable resistance value to avoid speaker frequency response interactions;


Vout = Vin x R2 / (R2 + R1)

ICONOCLAST™ total CMA size mitigates appreciable calculated frequency response changes, and stopped at 9600 CMA (10 AWG). 

For most practical applications of 0 to 35 feet, 9600 CMA per polarity should work well to be resistively invisible to the speaker, or amplifier. We want the load to be the speaker, not the cable.

HOW we get to the approximate 9600 CMA per polarity is the hard question. For those that want the easy way out we have one, 1313A. If we want to see if we can DESIGN a better MEASURING cable let’s see what can be done with Belden technology.

In order to figure out what best to do, I looked at things that indicate what NOT to do. We all know by now, that multiple smaller wires (to a point!) are better than one fat 9600 CMA solid or stranded wire.  The operative here is, to my ear, TIME based issues at audio. You want the signal to be most uniform through the wire for improved current coherence (more identical frequency arrival times). To make that happen, we decrease the wire size so that the skin depth penetration goes deeper into the wire, evening out the differences in current magnitude with respect to frequency.  This technique better aligns the signal speeds through the wire. I said “better” as there is no perfect way to do this. But we can certainly be better. The depth is calculated based on frequency and material. The wire size does not change the penetration, it DOES change the minimum current found in the center of the wire. The smaller the wire, the closer the center current magnitude matches the surface current as signal frequencies go up.

Studies were made on various geometries that would hint at what type of conductor to use, and how many. What various design limitations be “inside” the ~9600 CMA resistive box we want to be within?

Probably the easiest approximation for a cable with multi-sized wires is a flat design.  Yep, line those wires up and stop when you reach the proper AWG size.  The parallel wire tested issues lead (pun there?) me away from this simple design. Why? I looked at our TEFLON® ribbon cable for that answer.

A really nice “flat” TEFLON® Ribbon Cable

Above is a TEFLON® ribbon cable I used to test polarity symmetry, and capacitive symmetry WITHIN each polarity.  The two tables below graph the capacitance from the outer edge wire to the opposite polarity, all opposite polarity wires grounded together.

1 kHz10 kHz
STD DEV=0.8620.761
@1 kHz

The data says that the CONSISTENCY of FLAT cable is not perfect. The closer each wire gets to the opposite polarity, the higher the capacitance. The GROUND reference is more robust the closer we get, and the less distance between two wires, all else the same, the higher the capacitance. We have EACH and every wire, for all intents and purposes, acting like a different wire. ANY cable with more than ONE wire per polarity will have this issue to contend with. How can we do better on capacitance control in each polarity?

For the answer to that we need to turn to inductance. When you separate the two polarities in a flat design, inductance is seemingly well controlled. Each parallel wire has current going in the exact same direction in each polarity half so the magnetic fields CANCEL one another. The closer to the inside polarity separation zone you go, the more the opposite polarity’s different current direction upsets the SYMMETRY of the inductive cancellation process. There is non-linearity through the “flat” polarity, too, but it is worse near the edges of each polarity where the “design” changes.

Two wires with the SAME current direction next to each other cancel some of the fields’ gauss density between them, and two wires next to each other with opposite polarities reinforce the magnetic field lines.

Below are two close proximity wires. Notice that the current direction “adds” between the wires with the magnetic field flux lines in the same “reinforcing” direction. If we FLIP the current direction of one of the wires, the currents cancel but now we have two of the same polarity to get the cancellation effect. This is the problem with ZIP cord. We can get low capacitance, but it is not practical to get the lowest inductance.

To prove a point, a single bonded pair used in ICONOCLAST measured by itself is 12.5 pF/foot and 0.196 uH/foot inductance, about what 1313A reference zip cord is (chart below). This isn’t the best reactive variable balance of L and C for a premium current delivery cable.

In the tested flat design there are inconsistent ground plane issues that have to be resolved, AND there are inconsistent electromagnetic field cancellation properties, too, through the “flat”. The problems are locked-in by the geometry of this cable specimen, same as the issues with zip-cord.

What is GOOD about a flat cable that and can we use those positive attributes and mitigate the bad aspects? The answer to that question lies in a BONDED pair used at RF frequencies. To get to the answer for speaker cable, we need to re-invent what a BONDED pair does at audio. Re-designing a bonded pair for audio leads to what size and count wires we can manage in forward processes. We STILL don’t have the conductor size or quantity question answered after all this.

What is a bonded pair? A BONDED pair is two co-joined wires. A super geometrically consistent zip cord design with superior adjacent wire BOND technology. The precision C-C of each wire controls impedance at RF to incredibly small variation.


A zip cord removes a lot of symmetry complexity for poor magnetic field cancellation properties. Adding wires to the zip cord to make it a FLAT cable just adds to the capacitive and inductive “cable in a cable” issue as every wire becomes its own drummer. Coherence is improved with more small wires that add to the same CMA, but we don’t really have “one” like polarity for each signal anymore.

Tests show the inconsistent capacitance in a FLAT arrangement. Tests can also show the INDUCTANCE issues with zip- cords. A single bonded pair is 0.196 uH/foot inductance. This value is far too high for the state of the art R, L and C cable that is the intent of the project.

How is using another bonded pair zip cord component going to fix this mess? The answer is in the XLR cable. We need to build STAR QUAD arrangements of BONDED pairs! Visualize the currents using the right hand rule;

Like the XLR, two BONDED pairs in a QUAD arrangement show ideal field cancellation with LIKE polarity current all in the same within the same polarity. This field cancellation property of star quads tells us fundamentally we need two polarities using many wires in a star quad arrangement. There isn’t an answer as to how, yet, just that a true star quad is a key element we need to keep.

The solution was a compromise, as is usually the case in audio cables. The design devised a way to create star quads THROUGHOUT a process that varied between near perfect, and slightly imperfect. It was done with 100% consistency within each polarity so every wire measured the same inductance and capacitance to the opposite polarity, and made significantly lowered inductance with only a moderate rise in capacitance. The capacitance was increased on purpose, I might add! More on why I did that later.


The above illustration shows the variation in the STAR QUADS between like bonded pairs in a polarity. The question is does it work; capacitance measured 45 pF/foot between polarity wires and inductance measured 0.08 uH/foot. Capacitance variation, and the electromagnetically tied inductance variation, is superb.


The difference in reactive stability between each wire in a single polarity, and BETWEEN each polarity can measures significantly better in ICONOCLAST.

1 kHz10 kHz
STD DEV0.1660.202

Tolerance is +/- 0.5 pF @ 1 KHz or more than 5 times tighter variation than the 8R28064 flat cable.

What was done was to BRAID, on a GHz capable braider, the needed wires to arrive at the 9600 CMA DCR requirement. The braider needed a symmetrical arrangement so an even number of bobbins was chosen, 12.  This is 24 wires per polarity. 9600 CMA / 24 = 400CMA per wire, or a 0.020” 24 AWG wire.

The braid DESIGN is not forthcoming, so the balance of electricals has to be understood.  Several, several design iterations were trialed before I froze the design around the proper braid relationship to arrive at a suitably balance reactive cable measurement.


People will “guess” that ICONOCLAST is a BONDED pair ETHERNET cable, and it is not. The REASONS and the DESIGN are not the same at all. All that is the same is the coincidence of a 24 AWG solid copper wire common to Ethernet.

Each polarity is BRAIDED and FLATTENED into a, you guesses it, FLAT shape! We essentially “fold” the flat cable over on itself into ONE polarity. Then, opposite polarities are tightly bound to keep LOOP area to a minimum, critical to inductance as the formula is GEOMETRY controlled, not the dielectric.


Measured Rs (skin effect / proximity effects)

            The nature of the magnetic fields can be indirectly MEASURED with an Rs measurement. The flatter the Rs, the better the skin depth / proximity effect are managed. Proximity effect is the currents in each polarity being “pulled” to the inside edge of each conductor, and away from the outside edge. This impacts conductor efficiency.


An awful lot of testing was done to identify the weaknesses of various designs. We wanted to avoid;

  • Inconsistent capacitance in each wire.
  • Inconsistent inductance in each wire.
  • Inconsistent ground plane interaction between wires and between polarities.
  • Inconsistent wire DCR between all wires.
  • Poor polarity DCR values (too high or low total CMA).
  • Inconsistent dielectric performance between each wire.
  • Poor frequency coherence in each wire.

After all the testing, a 20-mil wire diameter in a 24 wire (12 bonded pairs) woven polarity was created to match the design to the electromagnetic requirements.  The final design that drove the final wire size is 100% symmetrical in every measure on every wire.

Woven single polarities achieve class leading performance in polarity-to-polarity and wire-to-wire consistency while also providing exceptionally low reactive variables. The superposition of the magnetic fields drive inductance down from 0.196 uH/foot to 0.08 uH/foot, a 59% reduction in inductance, while holding capacitance to just 45 pF/foot. L and C can be CHANGED based on the woven DESIGN, but was optimized for speaker cable applications.

  • Dielectric material(s).

TEFLON® was chosen as it is again, the best solid dielectric there is. I needed a thin wall to bring the wires close together for inductance reduction but capacitance is an issue with 24 closely spaced wires. A capacitor is two parallel conductive plates with an insulator between them. To lower capacitance, I wanted a low dielectric constant plastic, Teflon®. To achieve the required low capacitance, more needs to be done to “thicken” the insulation without increasing loop area effects.

This seems impossible to do, but it isn’t with the woven design described above. The final insulation wall was driven by BALANCING capacitive gains with inductive reduction.  Dielectric geometry allowed this balance to be accomplished.

  • Dielectric geometry.

The requirement to meet capacitance ALSO drove the design to a weave pattern. Each polarity is SEPARATE from one another. There is NO interweaving of same polarity wires.

Some will ask about wires with several AWG sizes. Current will flow along the path of least resistance. This does not mean current won’t flow in specific wires, just that the majority of the current magnitude is shifted to the easier path. EVERY wire will have current at ALL frequencies. The magnitude will change and follow ohm’s law. Many differing wires sizes and electrical lengths can impact the signal arrival times across the audio band based on physical conductor lengths in composite wire size designs.

If we take two wires with the same exact skin depth (same frequency point being considered) but one wire has twice the surface area, more current will flow into the larger surface area wire. It offers less resistance. But, the lower resistance wire is a larger wire and isn’t what we would like if the current across the wire is to be more uniform. Bigger wires are better at lowering resistance at a given frequency because they have the most surface area. We use this at RF with a “skin” of copper to carry the lowest, yet still high, frequencies efficiently. The wire’s core under the copper is a material that is “filler” and has no current flow: steel, aluminum, etc.

At lower frequencies the current is diffusion coupled evenly through the ENTIRE wire. So if you send JUST low frequencies, use low a DCR wire as you can get.

Those are the extremes. Audio is weird in that we need to improve current coherence through the wire while it is trying to MOVE to the outside surface. We don’t care about attenuation as much at audio since it is negligible. We make the conscious decision to go for forced current coherence with more SMALL wires. This technically violates the practice of more “surface” area for lower attenuation at high frequencies for current coherence. Big wire is more surface area for attenuation while small wire is better current coherence but higher attenuation.  If you use one wire (interconnect) the current delivery has to be considered to the load. RCA and XLR cables have near zero current flow into the high impedance load so we can go for signal current coherence and suffer little attenuation. Speaker cables can’t use too few wires as there are 20-30 amps coursing through a speaker cable.

Audio is trying to TIME align the low and high frequencies, so the best, and most consistent, way to do this is to use more small wires that add-up to the low frequency DCR needs, and are small enough to FORCE the wire to see more and more cross sectional current usage at higher frequencies.  This means several small insulated wire that all need to be the same “single” wire.

The unique woven design does a LOT to reduce inductance and associated capacitance. How is 59% reduced inductance over a single bonded pair achieved?

    • Star quad wire arrangement.
      • Allows ideal geometry for low field strength.
    • Boned pair like polarity wires.
      • Allows star quads to be formed throughout the weave.
    • Separate polarity halve fields are NOT parallel, reduce field reinforcements.
      • Fields between polarities have some cancellation (wires that cross at ninety degrees cancel) since the cross at ANGLES, and not ever parallel.
  • Controlled Proximity effects / Skin effects
  • Measured Rs flat to 20 KHz.
    • Low dielectric constant plastic.
      • Thinnest possible C-C with the lowest cap.
    • Woven pattern averages out the wire-to-wire distances significantly.
      • Woven pattern separates the wires and “tricks” the bulk capacitive value to be far lower.

The last point on the capacitive reduction is also what we like in a FLAT design, but it is inconsistent. Average distance between any two wires in a braided polarity and thus between polarities is far more consistent.  The weave moves all the wires evenly, and consistently, to a closest proximity position and a max proximity position throughout the weave.  Capacitance and inductance DO vary, but they are exactly the “same” wire and at the same time as every other through the weave. The fattened weave holds overall capacitance to an unexpectedly low value of 45 pF/foot in a cable with such high conductor count. 

Low inductance leverages the same current direction in the bonded pair’s combined with the star quad wire geometry periodicity (end view photo above). And finally, the TIGHT textile weave between polarity halves force a low loop area and with wires never being parallel, further reducing inductance.

The overall reactance of the cable is shown in the graph below.

The chart illustrates a significant drop (yellow trace) in cable impedance compared to 1313A (blue trace).  We know all we need to know to figure out why this happened. The velocity, although variable, is nearly the same at each SPECIFIC swept frequency point. We need to look at frequency by frequency calculations. The capacitance is linear across the entire audio band so that’s a set value.

We have a set value of capacitance, and a nearly set value of velocity (there will be slight variation) at a given frequency. What is CHANGING is fundamentally the capacitance between cable designs for “impedance” characterization.

The impedance equation is influenced by the change in capacitance and thus lower measured impedance as the capacitance shows up in the denominator of the impedance equation.  Increasing capacitance from ~16 pF/foot to ~45 pF/foot decreases ICONOCLAST cable impedance. Speaker cables require low inductance and to get there without shooting capacitance through the roof. DESIGN is the overriding requirement, and materials alongside unprovable theory, are second.

Now we know why ICONOCLAST has the capacitance it does, as I can balance the inductance to industry leading values AND keep cap low, yet not so low as to increase impedance too high relative to the input requirement (impossibly low speaker impedance 8-ohms ideal). Cables go UP in impedance as you drop in frequency, the opposite of what we want. Listening test have to decide if the superb inductance or impedance matching with much higher cable capacitance is ideal.  Quick calculations will show capacitance problems with 8 ohm cables at audio once an amplifier is attached.

Don’t ignore the reactive time constants of L and C. We want an 8-ohm cable with NO L and C and zero resistance and you can’t do that. Getting cable “impedance” reasonably low is more reliably safe for amplifiers and TIME based distortions (lower L and C).

  • Shield material and design considerations.

I kept this topic here on purpose.  Some may already know that low impedance cables signal levels negate the need for a shield. And that’s a good thing because a shield over a speaker cable is darn near ALWAYS a bad thing for two reasons;

  • A shield will always increase capacitance of the cable. The question is how much.
  • To mitigate the capacitance increase, the shield must be moved significantly AWAY from the core polarities, increasing the size of the cable.

Shields are ONLY beneficial if the environment demands them. Shields inhibit the performance of cable in most cases. Coaxial cables being an exception as the shield defines the cable’s natural IMPEDANCE.  The ground plane proximity and uniformity are vitally important with short wavelength RF cables. Coaxial cables do just that. Audio is not RF, and these shields are more FUD devices than actual benefits, especially in speaker cables that have signals orders of magnitude over the background noise. Incidentally, the woven pattern in ICONOCLAST has a built-in immunity to RF not that that RF immunity is evident in the use of the cable.

View a SHIELD as a rain coat; great if you have water flying around but a major hindrance if you don’t. Audio seldom needs shielding on low impedance cables and here is why;

Magnetic fields decay rapidly with distance; ratio of 1/x^3. The best defense is to MOVE the low frequency electromagnetic cables away from one another. The foil and even braid shields are higher frequency shields that are ineffective at much below 1 MHz.  Magnetic fields lines need low permeability shield material (something a magnet will stick to) to route flux lines away from sensitive devices. A faraday cage is an example you can put something into to do this. Low permeability metallic shields are a pain to use (stiff and heavy). DISTANCE is the best remedy.

For EMI and RFI, the foil and braid shields used on Interconnect cable will be fine for RFI ELECTRIC field issues, but NOT 20Hz-20KHz magnetic fields.  Interconnect cables MAY have wide band input op-amps that can be needlessly hampered by RFI on the line. Speaker cable signal levels are many, many orders of magnitude above the RF and ICONOCLAST speaker cables aren’t a good RF conductor due to the weave pattern in the design.

  • Jacket design and material considerations.

All ICONOCLAST cables use FEP as the jacket to reduce UV sensitivity, plasticizer migration and chemical resistance.  The cables are designed to last decades.

SUMMARY – Little has been left to chance in the design of ICONOCLAST cable.  All the products are born from strict measurements and the management of known electrical parameters. Belden’s philosophy is to make as low and R, L and C cables as technically capable. The improvement to some may be unimportant. To others, and using different systems, they can be significant. The closer we manage the knowns, the better the tertiary elements will move along with those improvements. All cables “react” differently. ICONOCLAST is designed to offer the most benign interaction possible between your amplifier and speaker by leveraging high speed digital design principles to the much more complex audio band.


I have four 1 meter XLR sets going from the bottom component, up to the pre amplifier component. The natural LOOP needs to be allowed to form, unobstructed from the rear wall. This takes about 12″-13″ of clearance from the back of the unit to the wall. The rack may be closer to the wall than 13″ based on the depth of the rack. If the rack shelf is 24″, and the device is 19″, the rack will be only 13″minus 5″ or 8″from the rear wall, for instance.

Space the racks and devices APART helps, too, so a 180 degree loop back for one device on top of another is too severe. A bend radius of 8″ or so is needed (see the picture). If devices are such that the XLR are well offset, a piggy-back stack can work. The distance BETWEEN the XLR jacks is what is important, up and down or left and right or a combination of the two works.

The XLR will twist no more than 180 degrees to plug in for ANY wire path. With a loop-back, a change in the vertical alignment, to the left or right, will force a TWIST in the cable. Same with a straight through if the equipment is shifted left or right. Again, this is simply how XLR works.

Plug the MALE XLR in first. The female has a thumb tab that lets you quickly see the direction to go that puts the least torque into the cable. Simply rotate the female XLR to snap it in. All polarized plugs work this way. The BAV is designed to simply have a very low bending moment design that absorbs the exact same twist, but with less force…thus it can take severe routing.

ICONOCLAST used with adequate rear clearance can use 1 meter cables, but the FULL natural loop needs to form to allow the torque to be absorbed along the full length of the cable. This is worst case as the jacks are pretty much one straight above the others.

ICONOCLAST are true AIR tube with the best dielectrics available. These cables will require proper routing and rack positions. For those that can not conform to the reality of the design, I have engineered the BAV to be the best flexing (first requirement) and best performing for the flexibility. These will accept harsh routing.

Galen Gareis

iconoclast design engineer


If you have spent plenty on cables you may well wonder WHY these cables are physically as they are. If care is taken to adhere to fundamentals, there are very good reasons for a physical design in audio cable, of both high (interconnect) and low (speaker) input impedance types. If we look at all the fundamental electricals through the audio band, is it any wonder every cable doesn’t sound different? Let’s see why that might be, and no magic need apply throughout this analysis.

What is happening in audio frequency ranges?

  1. What exactly are we “moving” with zero distortion?
  2. Current and Phase Relationships.

3.0 Electromagnetic wave propagation differences with respect to frequency.

4.0 Impedance and matching to a load at audio.

5.0 Capacitance and Inductance with respect to frequency.

6.0 Cable Capacitive and Inductive reactance properties rise and decay time distortions.

7.0 Current normalization / skin effect.

8.0 Dielectric effects.

9.0 AC resistance changes and frequency.

10.0 Cable symmetry issues.

11.0 Attenuation at audio.

12.0 Passive low pass filter effects.

If we look at pure tones; sinewaves, square waves, frequency and TIME are interchangeable. Math says that this is so, and there isn’t anything new that explains that away. When we add TIME based distortion to the sound delivery system our ears are quick to “hear” the deterioration in fidelity based on frequency arrival time and phase coherence more than amplitude limitations (attenuation).  How much is a cable responsible for this?  The superposition of the 12 listed distortions (and there are more) are much more significant than any one taken on its own. There is truth to the concept that slew rates, or how fast a system responds (wider bandwidths), affect performance. A square wave is but a multiplicity of sine waves. Mathematically every frequency’s characteristics, at every point in a cable can be predicted.  Cable is far from perfect at moving electromagnetic wave through the audio band, however well we can calculate the accumulating TIME based distortion as the electromagnetic wave moves down the cable. Better designs minimize those distortions and place more or less emphasis on each one depending on the designer engineer’s concept of audible influences. The fact remains, cable design is still driven by the DESIGN needed to reach the R, L and C values with minimal influences on tertiary elements. Can you hear a more fully optimized design? This is why we present these designs for audition.


The issue – What do we actually LISTEN to on a cable? What is the “root” reason to be for a cable?

Cables exist to move the “signal” from one place to another, but few really consider WHAT that signal is. The signal we “use” is the electromagnetic wave moving down the cable at the group velocity of propagation of the dielectric. OK, what did I just say? Imagine our wire surrounded by a donut with a hole in the middle! The electromagnetic wave is this donut. There is an ELECTRIC (E-field) around our wire too, but this field is attached to the donut radially, and ninety degrees orthogonally to the donut’s circumference. To make the E-field, take a bunch of tooth picks and stick them all around the outside of the donut, that’s the E-field.

Now we have two imaginary waves, one low frequency and one high, sitting there. To MOVE that field, electron flow starts it happening.  To keep it simple let’s distort our wire to be a TUBE full of marbles (electrons) that has an inside diameter the same as the marble’s diameter. To make the magnetic field move, and drag along the E-field with it, we apply an electromotive force (electrons / marbles) to the tube. When a marble is inserted into the end of the tube, the marble at the opposite end pops out as fast as the marble can be inserted into the send end of the tube. This “speed” is determined by the velocity of propagation of the dielectric, or the tube in our case. Something funny happens with the magnetic field though; it follows the PROGRESSION of the electron (marble) flow. When the marble is half way into the send end of the tube, our donut with all our toothpicks (the B and E fields) is halfway down the cable already!  When the marble is inserted all the way in at the send end, the B and E fields are at the END of the cable. So the “signal” we use travels at the VP (velocity of propagation) of the cable, and NOT the speed of the electrons at all. Those move very slowly compared to the electromagnetic B and E fields. Now we have the donut at the end of the cable. But, we won’t ever see a baker’s dozen, or zillions more moving electrons appear at the same time at the opposite end of the cable if we carry more than one frequency concurrently since every frequency has a different VP through the audio band.  All individual frequencies will have significant arrival time “distortion” between frequencies. In other words, every marble that represents a frequency in my example is inserted at a different speed (Velocity of Propagation) depending on the frequency the marble represents. Ideal cable should move a signal (now we know it is the B and E fields) down a wire at the same speed and shape at all frequencies. It doesn’t.  

2.0 Voltage and Current Phase

The issue – Current and voltage are locked into a phase shifted relationship, always.

The reactive properties of inductance and capacitance are responsible for a ninety degree time based shift in all electronics, not just cable. There is a common ditty about the current to voltage phase relationship that goes like this; “ELI the ICE man”. It is a memory tool to remember that voltage (E) leads current (I) in an inductor (L) and that current (I) leads voltage (V) in a capacitor (C).

Why is this? A capacitor has to charge with applied current to reach a steady state voltage, so as the voltage potential increases the current drops. The current has to be there BEFORE the voltage potential hence current leads voltage in a capacitor.

An inductor resists current change when voltage is applied. Current reaches a steady state over TIME with applied voltage, so as the current potential increases the voltage drops. The voltage has to be there BEFORE the current potential hence voltage leads current in an inductor.

These two locked-in relationships lead to all sorts of other TIME based issues in cable and circuits. They are the variables that constitute PHASE in an impedance trace, for instance, and reactive TIME CONSTANTS that we’ll cover later in the paper.


The issue – VP varies the arrival time of signals moving down a cable. Signals should ideally leave and arrive at the same time and shape as they are sent at all frequencies.

Audio is in an electromagnetic transition band. This is the elephant in the room. It prevents cable from EVER being perfectly accurate when moving low frequency electromagnetic waves. The propagation constant, the speed at which the electromagnetic wave / signal moves down the wire’s outer circumference, and not IN the wire, is determined by the dielectric material that the electromagnetic wave is predominantly traveling through. We can measure this effect directly and indirectly.

At RF, where life is way more consistent for cables, we can calculate the velocity from the DELAY equation. For Ethernet cables the following equation is used;


The delay equation uses FREQUENCY. This is a TIME based value so it tells us that we have arrival time issues as the frequency changes, and less so at RF, and WAY more so at audio frequencies. The table illustrates the slow erosion of speed as we reduce the RF frequency. A little change is evident but audio frequencies see much more change.

Delay values measured at RF (MHz)

Actual data shows what audio cables do; the impedance RISES as we go LOWER in frequency, by a lot. This is because the DELAY / VP factor drops, and adds TIMING issues to signal delivery.

Above are actual traces of how ICONOCLAST performs across the audio frequency band vs. typical zip cord speaker wire (1313A) and out to RF, to prove a point. The impedance increases considerably below the RF frequency reference values. Those 87% and 90% VP factors we love to “hear”, high VP, are clearly not valid in the audio band.

How significant is the VP change? In the example above we drop from ~110,000,000 m/Sec @ 20 KHz to ~5,000,000 m/sec @ 20 Hz or a factor of 22 times slower through the audio band.

To make matters worse, it is a LOG function so it is not linear. This is what physics has thrown into the design process.  Can we hear this change? Attenuation at audio is a passive linear variable and considered to be insignificant (keep your cables short) but every variable keeps adding up to the overall actual performance.

Notice that the cable’s impedance, made for audio not RF, flattens out @ ~ 50 ohms above 100,000 Hz (see the table below for the actual values). Just because something has an “impedance” (real and reactive L and C component) does not mean it is a transmission line.

Look at the low-frequency range. Isn’t cable supposed to be the same at all frequencies or the same TIME base? The velocity constant at a frequency is TIME, so the fact that we see a difference indicates a non- linearity across the usable audio band. The problem is that thing called propagation velocity (VP) or the speed that information travels at differing frequencies in the cable.

The equation at audio compared to RF is more complex (wouldn’t you know it!);

Z = sqrt((R+j*2*pi*f*L)/(G+j*2*pi*f*C))

impedance (Z),

capacitance (C)

inductance (L)

resistance  (R )

conductance (G )

Using the general simplified RF equation, where all the extra stuff in the complicated impedance equation at audio goes to a one or a zero and drops out, we are left with; 101670 / (Capacitance x Velocity) = Impedance. At RF for ICONOCLAST speaker cable;

101670/(VP*45pF/ft) = 50 ohms @ RF

     Solving for velocity of propagation we see it is no higher than 45% at RF. This isn’t RF cable, and the design changes necessary for audio are what ICONOCLAST is after.  We need to ideally FLATTEN the VP curve for audio cables to better time align the signal in the frequency range where we use it.

The calculated graphs using a 75-ohm coaxial cable below show that VP change as we go lower and lower in frequency. Look at the IMPEDANCE at audio frequencies shoot way up, and the VP drop like a rock in a pond. Notice, too, that VP begins to flatten out at 100,000 Hz, just like the charts above on ICONOCLAST. This is real stuff, and it won’t go away…you have to MANAGE it to a balance in each cable.

What does our measured data show that corresponds to the theoretical chart above? Below we see several BELDEN products measured VP drop considerably from RF to, and through, the audio band. And, the measured values are near the exact same values I will calculate from measurement on ICONOCLAST; ~ 5% VP to 50% VP between 20Hz to 20 KHz.


The impedance goes up as we go lower in frequency because the velocity keeps going down, and the alternative variable, capacitance, just sits there (we’ll get to that soon). We have a differential in signal velocity across the audio band. Also notice that typical 1313A ZIP cord behaves much worse than ICONOCLAST™, rising to double the ICONOCLAST reference impedance value. Be warned, audio cable does NOT respond to impedance matching like RF.  

Speaker cables are theoretically designed to be much lower impedance, and terminate into reactive 2-16 ohm loads, and some point way north of 16 ohms.  Interconnect cable is terminated into “high” impedance resistive loads of 47K to 120K or higher, and should be much higher theoretical impedance than speaker cable, and the graphs above show exactly that.

It is good to see impedance matching to a load, but other variables are in play, and impedance matching isn’t meaningful or practical at these frequencies and impedances. Good designs usually address ALL parameters, however.

Interconnect and speaker cables, with VERY low audio range VP values show a much faster VP in the RF band. The values of 87% VP @ RF are NOT really correct for WHERE the cable is used, but “sounds” exciting.

What do we see at RF on an ICONOCLAST interconnect cable? We can calculate what we measured in the graphs above. We can use a grossly simplified equation to predict the VP based on capacitance measurements;

101670/(11.0 pF/ft * VP) = 105 ohms @ RF

Solving for VP we get a value of 88%, using the measured values of 1 KHz referenced capacitance. This VP factor will DROP considerably in the audio range to much LESS than that. Imaginary values (L and C) stay the same from 1KHz to RF frequencies so VP is changing;

101670/(11.0 pF/ft * VP)=~2000 ohms @ 100 Hz (audio)


VP = 4.3% @ 100 Hz. (101670 / 2156 Ohms * 11pF)

VP = 57% @ 20 KHz (101670/163 Ohms * 11pF)


VP = 2.17% @ 100 Hz 101670/ (278 Ohm * 45pF)

VP = 55% @ 20 KHz 101670/ (41 Ohm * 45pF)

If we take the VP reduction factor of a coaxial cable into the audio band @ 22 X lower, we see; 87% / 22 = 3.9% @ 100 Hz. Close to the same answer in our rough calculation.

The data shows a 13X to 20X or so DECREASE in cable speed as we drop in frequency. Signal arrival times are NOT staying in perfect symmetry relative to the input start point. The AMPLITUDE may be near the same, but the TIMING is certainly not. Arguments persist as to how long the cable needs to be to her the arrival time coherence.


The issue – All cables should terminate into their characteristic impedance (not really true at audio). At audio, the cable isn’t a fixed impedance, or even really an “impedance”. Interconnects see a resistive “infinite” load, but not speaker cables, which see a highly reactive low impedance load. 

Impedance is a REACTIVE vector value. This is a dead giveaway that we’ll have to deal with Dv/Dt stuff. All cables are a wire that is in series with an inductor and a capacitor to ground. All three R, L and C, keep getting bigger the longer the cable on a bulk value basis. The impedance is a VECTOR sum of the REAL part and the IMAGINARY part. The PHASE is created by the imaginary part of the impedance vector value. The impedance values aren’t the same for all frequencies (see the 1 KHz and 1000 KHz chart below) since VP keeps changing, and this is a component of the impedance value. Since the impedance is a vector sum magnitude ratio, it stays constant for each frequency point no matter how long the cable is. R, L and C increase proportionally.

      Reactive Change with Frequency

Most of us kind of know that we are supposed to match the impedance to the load for the best transfer of energy. We are actually only terminating the resistive component we call “impedance” to the load; a resistor in the case of interconnects, or a speaker load for low-impedance speaker cables.  There is a reactive component that is also at issue for good signal transfer. That reactive (usually capacitive) part of the Impedance vector magnitude diminishes the transfer of energy in time. Audio is not RF, so this matched resistor to resistor ideal isn’t exactly correct anymore, even for high impedance interconnects. The physics of the velocity of propagation make impedance matching impossible at audio as does the wavelength, which is far, far too long to react like a true “impedance” vector.

For transmission line effects to be a factor, the cable length also has to be at least 10X or more the quarter wave length of the frequency of interest. This relates to the fact that a voltage change has to happen BEFORE it gets to the end of the cable and audio speaker cables transit times are too fast, even @ 50% VP, for this to happen.

A cable can have impedance (real and imaginary values), but it is largely irrelevant to true load matching. There can be a signal reflection based on the CUT length of the cable relative to the speaker. This simple reflection can be absorbed with a ZOBEL network across the speaker terminals if it induces amplifier oscillations. But, low cap cables are benign to amplifiers, even with this simple length defined reflection.  The cable will sound the same with or without the network as the parallel circuit is not in the signal path. The tertiary effect of better amplifier stability is what improves the sound with too high capacitance cable.

At RF, a signal is “used” efficiently only when two like resistive loads see each other. RF cables are designed so that the cable impedance matches the restive termination load. Audio cables don’t work like this at such low frequencies since we can never transmission-line “impedance” match to a load with short passive cables. But, the “work” done across the load STILL has to be resistive. The imaginary components of a vector (Impedance is a vector sum of the real and imaginary components) store and release energy since they are composed of reactive variables; Capacitance and Inductance, both variables, are store and release variables of voltage and current respectively. Short cables still have reactance.

We can see what happens at RF. The graph below shows actual cable data of what is called Return Loss. The return loss, RL, represents the “reflected” signal that does not transfer to the load for an RF Ethernet cable. RL= the imaginary part that can’t do work till it is “real” or resistive. Notice that we see several RL values “dead nuts” on 100-ohms from a low of -55 dB to a high of ~ -22dB. WHY are the RL variables not all the same? The impedance shows 100-ohms for all those RL values. The impedance at every frequency has a different reactance due to a lot of things too complicated to explain today. Simply put, at the frequencies with the lowest imaginary component, more energy is transferred to the load. In our example, if the impedance is above or below 100 ohm, and more or less reactive, the RL is decidedly worse. This is the cause of the FAN shaped graph that we see below.

Audio cables aren’t used at RF, though, and suffer from simple reflections more than load matching ones. This isn’t bad thing, as the critical attributes at RF aren’t restricting what we need to do in the audio band for better signal quality. We don’t need to worry about minute wire diameter fluctuations that cause the above graphed RL reflections. Audio wavelengths are too long to see the diameter variation issues so designers can work with geometries that may not be ideal at RF, but are far more useful for coherence adjustments in the audio band. Those adjustments still have to be real, of course, and measured or calculated with accepted standards.

Audio speaker cable with AC signals is terminated into a load that is resistive and reactive. Alternating current reacts to the imaginary circuit cable values and regulates how fast, and when, we can get work out of the cable. Some early cables were so reactive that amplifiers would shut off using them. Even though our cable is not a true impedance we do have reactive elements.

Interconnects see an “infinite” ideal resistive load; 47K-ohm on up, and speaker cables see a very low, and varying, reactive input impedance (the impedance of all loudspeakers changes with frequency).

Speaker cables are CURRENT signal devices that are designed to transfer power to an electromechanical motor. And, a motor that constantly “changes its spots” at every frequency as does the cable. The “argument” between the speaker EMF and cable is complex. 

Interconnect cables are VOLTAGE signal devices terminating into a HIGH impedance resistor. We want to transfer the signal shape and amplitude to a load. To avoid distortion(s) we don’t want the cable or the load to mess with the transmit circuit, but they do.

Audio cables are way too short to be transmission lines, needing at least 10X the wavelength inside the dielectric to be a true transmission line. Even 20 KHz is way too long a wave length to match that definition. We DO have simple reflections off the LOAD (speaker itself) that cable can’t manage as the load varies with frequency. This is very different than RF where I can make a cable nearly look like the load, minimizing reflections. I said “nearly” as all cables exhibit reactance, a TIME based storage of energy. Audio cables have significant measured time based propagation error due to VP and now we add-in a rise time error from reactance. The reactance of cable can be used to calculate “time constants”. At audio every frequency is associated with a different constant value. We’ll look at time constants later.

Zobel networks have been used to good effect to dampen the cable to speaker load variation, but they are estimations of where the two are most aggressively reflective. A Zobel network is a passive means to connect two differing but fixed characteristic impedance lines with a resistive value. Neither the cable nor the speaker are linear loads making it an approximation as to where to tune the Zobel network.

For more on Zobel networks and speakers go to;

Zobel networks and loudspeaker drivers

Compared to our “typical” Belden cable (blue trace), ICONOCLAST is flatter (orange trace) in velocity change as we go lower in the theoretical impedance. This is more the result of a higher, but still low, capacitance between the two designs. Lower inductance was preferred over capacitance.

The table data below is REAL and represent what even really good cables do through the audio band. The physics of the propagation delay match the measurements.


The interconnect tables follow and yes, they too show time based changes.


            The issue – What do the reactive variables do with respect to frequency?

Capacitance and inductance are essentially FLAT with frequency. Yep, capacitance and inductance are, interestingly, the same from near DC to the “sky is near the limit” frequencies.  Capacitance is set by the dielectric, assuming it is a linear dielectric material, and some aren’t (PVC). Measurements show that stable dielectrics offer frequency linear capacitance. Inductance is set by the distance between the wires and the loop area; it isn’t changed by the dielectric at all. These two values are always steady Eddies, but their time based effects on current and voltage change with frequency.

       Here is ICONOCLAST speaker cable that shows L and C data,

and it is FLAT with frequency using Teflon® as the dielectric.

The choice of what plastic to use sets the dielectric constant. You want stability with respect to frequency. Teflon® has the lowest dielectric of any SOLID plastic and thus the lowest capacitance with the thinnest walls of any material and, it is durable. It costs a LOT to buy and process, too.  Cost isn’t why we use it, performance is.

Plastics aren’t magic for capacitance, that is just the way it is. You want to pick the lowest dielectric constant value not just for low capacitance, but to help offset the change in the dielectric constant with respect to frequency. PVC dielectrics are far worse in linearity with respect to frequency, and the slope is not the same everywhere.  The chart and graph below assumes a set wall thickness and changes to the dielectric material alone. We can alter the WALL thickness based on the dielectric constant to get a given capacitance between two wires. Double the dielectric constant means doubling the wall for the same capacitance. Use the cheap stuff then? Sure, but more wall thickness increases loop area (space between the wires) which increases inductance! Oops, we’re not going to get zero cable reactance that way! A wire in a vacuum inside a braid ground would be the smallest size with lowest capacitance you can realistically see. This design would also have the lowest inductance since the loop area would be at a minimum with the vacuum acting as a low dielectric material.

We can calculate the effects of the dielectric and capacitance using a shorthand RF formula 101670 / C *V. We fixed the reactive impedance to a set value, so for a fixed wall of insulation, the capacitance rises as the dielectric constant is higher.  Since we know that the capacitance value is flat with frequency, this applies to the audio band as well. Better dielectrics for a given wall mean lower capacitance. This has nothing to do with Inductance, which is related to the magnetic field lines. Inductance is related to the distance between conductive surfaces, the less the better and field cancellation…if any.

Impedance = 100 ohms

Velocity = 1 / SQRT (E)

Capacitance = 101670 / (impedance * VP)


The issue – all cables store and release energy (current or voltage) reactively to the frequency being electromagnetically moved through the wire, adding time based distortion.

Look at the Impedance / Phase trace shown above on Part 3, Velocity of Propagation Issues. Notice that the PHASE on BOTH cables changes. PHASE include reactive components that TIME shift the signal’s ability of a signal to become resistive (If the phase trace hits “0” the circuit is resistive and has no reactive component).

A capacitor looks like an OPEN to DC or to very low frequency AC voltage changes. The cable is very “reactive” to voltage changes at lower frequencies. As you go up in frequency the cable’s bulk capacitance looks more and more like a SHORT circuit. The cable becomes more “resistive” looking with less reactance to voltage change. The trace explains why we can use 75-ohms and 100-ohm loads for RF cables, they look “mostly” resistive at RF.

Using a high impedance probe to measure the cable’s reactance produces the traces that you see. There is little current flow into the cable and, this is essentially how interconnects are used. The terminating load is VERY high impedance limiting current flow. When you put a voltage across a capacitor (our cable) it sends a momentary inrush of current to try to fill the capacitor. Output devices loading the circuit are ideally super low impedance to allow for this “inrush current”. Cables with lower capacitance mitigate the inrush current issue. Current LEADS voltage in a capacitor so there is a TIME shift caused by the cable.

Speaker cables are differing in that we don’t measure them like they are used; terminated into what is essentially a short circuit, the speaker. The large current flow in speaker cables responds to reductions in INDUCTANCE. Inductors resist current flow changes and that’s what speaker cables are trying to “move”. Voltage leads current in an inductive circuit and again, we see a TIME shift caused by cable but the opposite reactive variable, inductance verses capacitance, than the interconnect cable. 

Also consider that in speaker cables, the most reactive region is exactly where speaker’s impedance is also the most reactive, too. We want is a cable that is purely resistive but that’s impossible since a cable is a vector of capacitance and inductance.

Can we look at this another way? Yes, we can. If we examine the capacitive reactance equations below, and stick in the values at DC (F=0) and infinity frequency (remove F) and see what the results are we get the same answer; reactance is high at low frequencies and lower as you go up in frequency.

Xc= ½ * pi * F * C

XL = 2 * pi * F * L

The inductive reactance is the opposite, it looks much smaller at DC (F=0) than at higher frequencies (F= infinity). An inductor is a SHORT at low frequencies and an OPEN at higher frequencies. Fortunately speaker cables are relatively lower frequency making things less severe than at RF.

Cables, and all circuits, have capacitive and inductive reactance. Capacitive reactance resists voltage change and inductive reactance resists current change. They are both frequency dependent.

The TIME it takes to CHANGE the signal applied against a reactive load is measured in TIME CONSTANTS. It takes about 5 to 6 time constants to reach steady state amplitude. Our signal is also distorted the longer it takes to reach steady state amplitude so it may get nearly as big (we’ll pretend attenuation isn’t an issue), but it isn’t the same SHAPE. Don’t forget, every frequency is associated with a different time constant, and the decay or removal of the signal is the inverse. It takes TIME for the signal to bleed away to zero and this alters the decay signal.

As frequency changes, so do the reactive variables the determine a cable’s reactive performance.

At the very high end of the graph below, we see simply SQRT (L/C). At the low end the simple reactance (denominator) enter in.


            The issue – Current magnitude normalization at audio frequencies. Is this real?

There are several ways to calculate skin depth, and they all will yield the same answer. Impedance / RL can be derived from several inter-related factors and so can skin depth. It is real, and it can be managed to control phase distortion.

We all know about skin effect, but WHAT exactly is it doing at audio frequencies and is it real? Yes, skin effect is real at audio and all industry accepted calculations show that it is. The definition of skin depth is the point inside a wire where the current decreases to 37% the surface current magnitude. Skin depth is always the same depth of penetration no matter the wire size. Skin depth will vary based on the material’s electromagnetic properties and the frequency of the signal. For audio we calculate @ 20 KHz.

µ = permeability (4π* 10-7 H/m), note: H = henries = Ω*s

π = pi

δs = skin depth (m)

ρ = resistivity (Ω*m)

ω = radian frequency = 2π*f (Hz)

σ = conductivity (mho/m),  note: mho [Electrical ‘mho’ symbol – RF Cafe] = Siemen [S]

At low frequencies it simplifies to;

δs  = SQRT (2ρ/ ωu)

Looking at COPPER, we would calculate 461um (0.0181” depth).

If the skin depth at a given frequency is 10-mil on a 100-mil wire the 37% current point is well near the wire’s surface, it’s just 10-mil away in 50-mil radius.  If we halve the wire size, the current magnitude is larger through more and more of the wire. Each time we decrease the wire size, the larger the current magnitude becomes across the wire relative to surface current. In our 18-mil skin depth wire example above, the current in the “center” of a 36-mil wire will see 37% the magnitude of the surface current. Making wire smaller will INCREASE the current magnitude in the wire’s center to be closer and closer to the surface current in the wire at higher frequencies.

AC resistance involves FREQUENCY which is a TIME based variable.

RAC= (RDC) (k) SQRT (Freq)

K is a wire gauge factor that involves skin depth.

Freq is in MHz.

The internal wire impedance (AC resistance) is driven by the INTERNAL magnetic field’s relationship to inductance. Inductors RESIST instantaneous current flow and have higher “resistance” as AC frequency goes up. Current flows in the least resistive part of the wire as frequency goes up, so it reaches the surface where the self-wire inductance is nearest to zero.

Once we flatten the velocity change as best we can with a good dielectric design, we need to ALSO time align the effects of the dielectric at ALL frequencies using SMALL wires. Small wire improves arrival times as it forces the effects of the composite dielectric speed to be more uniform, as best we can, at all frequencies. This counters the skin effect problem that moves the current density magnitude to the surface of the wire as frequency goes up. Smaller wire increases the current magnitude (arrow length) in the wire center region to make it more efficient at time alignment.

One BIG wire
More SMALL wires

Even if we have the SAME current magnitude throughout the wire at all frequencies (impossible unless our wire is one atom in size) the velocity of propagation of the electromagnetic wave energy is STILL different at every frequency! But the ears say if we MANAGE the problems, our cables can sound much better. I took the time to measure all of this and flattened the impedance trace as much as I could. The VP changes less with frequency the flatter the impedance curve. Capacitance stays the same at all frequencies, so this VP is therefore changing less the more consistent.  Smaller wires are more consistent dielectrically at all frequencies.

Bigger wires will cause even more signal speed change relative to frequency because each electron’s is far smaller inside the wire. Each magnetic field contribution changes velocity the closer or farther that electron is away from the dielectric material. When a current is applied (electrons start moving) an inner wire located high frequency current mode travels slower than the same frequency signal on the outer wire surface and all these current modes are superimposed one on top of the other. This is called group DELAY.

Not all signals at the same frequency arrive at the same time, it depends on WHERE they traveled (MODE path) through the wire and  what the velocity of propagation is from the geometric perspective. The lower in frequency you go the less you can change the group delay since the current density through the wire is more and more consistent.

The overall magnetic field is a summation and superposition of ALL the moving electrons, the whole “group”. This is also why air is often used in interconnecting cables to mitigate the dielectric’s impact on the signal, and why you see more small wires in speaker cables. Electromagnetic field uniformity in the dielectric is important. The overall audible improvements are more debated. But, there is science involved in the optimization process.


The issue – dielectrics can impact weak electromagnetic signals disproportionately. Electromagnetic fields are squared law fields, and are most influenced by dielectrics nearest the wire. Weaker electromagnetic fields are most susceptible to dielectric distortions and the group velocity is mostly set by the strongest signals dielectric medium.

Using too many small wires splits up the current and starts to allow the dielectric to influence the sound more and more, negating the “advantage” of dielectric uniformity. The electromagnetic field is strongest nearest the wire, decreasing with the square of the distance moving out away from the wire. The electromagnetic signal moves from being “in” the dielectrics to being around it. The signal propagation speed is an average of ALL the dielectrics, with the material the stronger fields reside in have the most influence on the average of the “group”.

Four-fifths or more of the current magnitude at audio is below 3 KHz. Some call this the spectral power density, or roughly where the most energy is being placed. The electromagnetic energy does not STOP in the plastic or air. It emanates out in an inverse LOG power decay THROUGH all the materials it encounters along the way. The predominant material VP effect occurs CLOSEST to the wire. The smaller the signal (interconnect cables) the bigger the effect of the immediate dielectric nearest the wire.

Weaker signals will be impacted by the dielectric’s effects more than stronger ones, as they decay to far weaker signals moving away from the wire. The speed is more and more determined by the dielectric near the wire as we go up in frequency. Interconnects see little of the plastic out away from the bare wires as the field decays so quickly, but, the smaller the electromagnetic signal, the MORE it is influenced by the material nearest the wire. That superposition of materials SLOWS the signal (air to plastic) or speeds it up (plastic to air) relative to just the initial material’s properties.

We can see this in actual practice as the “group” velocity of all the materials on Ethernet cable shoes a value SLIGHTLY higher than the dielectric (66%) itself, and measures 71%. The signal is in the “air”, a good dielectric” and this influences the overall signal speed.

ICONOCLAST interconnect design switches this around, and puts the AIR nearest the wire, where the signal strength is highest. This negates the outer plastic dielectric’s contribution to the group velocity, so we see a higher 87% value at RF. This translates to lower capacitance number where we use the cable in audio applications.

The VP speed variation caused by the “composite” velocity is complicated by the fact that the LOWER in frequency you measure, Mother Nature’s devious plan slows everything and this time shifts the audio band.

We can’t change the fully diffusion coupled (same magnitude current through the wire) low frequencies, so we try to time align the faster upper frequencies. At RF the upper frequencies are “on” the wire surface so the dielectrics affect them nearly 100%. At RF this is fine because it is near the same VP at all RF frequencies. At audio, we want to move most of the high frequencies AWAY from the dielectric so the speed is closest to the lower frequencies.  We already know that the VP is faster the higher in frequency we go so this messes up the signal arrival times.  The only good way to slow the upper frequency magnetic field is to make the wire smaller so less energy is JUST at the wire surface nearest the dielectric. More current is “in” the wire versus “on” the wire based on skin depth.

Can we overdo field current normalization? What if we could make a wire one atom wide? Now, the impact of the DIELECTRIC is as big as it will ever be and with a really, really small current in each wire. The total current will be the sum of all the wires we want to use in parallel. The more wire you use, the smaller the current in each wire. Current is the number of electrons past a point with respect to time. Well, we have ONE tiny electron moving in each “wire” and THAT is as small a current as you can have! Model a weak signal, and the electromagnetic wave is so weak it never really leaves the dielectric, whatever material the dielectric is. The dielectric better be really decent as it is hugely involved in capacitive rise time (calculated capacitive reactance rise times constants) signal arrival time (velocity of propagation).

At very high frequencies, and if the wire is infinitely big, we see ONLY the dielectric as the current is at the surface (skin effect). Likewise if the wire is infinitely small we AGAIN see JUST the dielectric (no skin effect can happen). Between the extremes of wire size, somewhere, we can alter the arrival time of the upper frequencies with wire diameter and dielectric choices.

Interconnects are much easier, but not real easy, as they terminate into a high resistance, nearly open looking circuit. The reflections off a CONSISTENT resistive load of 47K-120Kohm aren’t as bad as the mismatch speaker cables experience as BOTH the cable AND the load are in constant flux. Worse, the speakers change by design! The seemingly high measured impedance slope of RCA or XLR interconnects in the initial graphs aren’t as bad as they seem. Not only are the “impedances” not real at audio but you have far bigger issues with the non-linearity of cables loading the output devices in your preamplifier. Trying to match ideal infinite input impedance on RCA or XLR cable would mean tiny capacitance values. We go as LOW in capacitance as we can to allow the output devices to see an easy load. This is way we shoot for keeping capacitance reasonably low.

Massive signals in the speaker cable are less impacted by the “composite” dielectric speed. The electromagnetic field will travel at an “average” of all the stuff it is moving through, so the better the “average” material is that the electromagnetic field is in, the FASTER the signal travels, and the less TIME the signals have to become separated as they travel down the cable. This is the time and distance story problem.

PC’s stopped using FLAT cables because the signal arrival TIME differential got to be too high. They went to SERIAL digital designs, and re-clock the data from memory. This at first seems counterintuitive, adding the re-clocking circuit, but unless the TIME can be managed, you’re screwed. Faster is better but I’d take SLOWER and the SAME in an instant! This is the “keep cable shorter” thing, but to be LONGER we have to be FASTER, too, if time errors are to be kept low. Mother Nature says we get a raw deal in the audio band verses RF.

In speaker cable, the stronger low frequency electromagnetic waves emanate into the air through the plastic dielectric more than the higher frequency signals so they are theoretically aided by the air around the wire (superimposed dielectric value) more than the weaker high frequencies that see more of the slower plastic dielectric. But the VP erosion as we drop in frequency eats-up that advantage in the low-end. It’s there, but small. The problem is that the low frequencies still drop in speed way more than the air’s addition to the overall speed. Seeing more air as we go lower in frequency speeds the signal up relative to the faster high frequencies and offsets some of the problem…but it never aligns it away to zero. The VP still marches slower and slower as we go lower and lower in infrequency. Arrival times are more important than SPEED down the wire.

The highest frequency carried in the speaker cable is most fragile, but compared to interconnect cable, it is relatively robust.  The high impedance interconnect cables are yet another problem. ALL the signals are VERY, VERY low current electromagnetic field energy states. Here, I need the BEST material possible to time align the energy field “whipping” (slowly whipping) down the wire; air. The VP is the inverse of the dielectric constant so we want a fast dielectric and the lowest associated capacitance it can also provide. This is why I HAVE to use AIR core designs to properly time align the energy AND use SMALL wires to better distribute the dielectric’s effects at ALL frequencies nearest to the same composite velocity. The third leg is to decrease output device capacitive loading. Air helps mitigates velocity variation across the frequency band that is the bane of audio signal transmission. It incidentally also pushes UP the “impedance” to better match the load, the opposite of a speaker cable. I’d be wary of that improvement as we need to be aware that audio isn’t a transmission line.

It seems counterintuitive to use air, as it speeds up the higher frequencies relative to the lower frequencies (makes the difference worse) but the capacitive reactance influences rise time error if you let it get too high. The propagation time and the rise time need to be balanced, somehow. There is no perfect solution.

We call, it “sound quality” when we use the cable, but it really is the arrival time alignment of all the signals. The human brain hears superimposed time alignment and amplitude preservation first, everything else a distant second. The argument is: does this make a difference?


The issue – Changes in electromagnetic properties between interconnect cables types can alter the ideal “tone” that was intended.

An often ignored issue is, what do you do with a really good sounding RCA cable? Why not make a really good sounding XLR that’s the same reactive measured design? Most RCA to XLR cables never match. ICONOCLAST is no accident. I purposefully designed the RCA and XLR to be the exact same reactive match and thus the same “quality” of sound through the channel. The above impedance chart that we saw earlier shows both the RCA and XLR. Look closely, they are electromagnetic buddies.

Does that make a difference? If you have a very good RCA design, it sure can’t hurt to start there on the XLR!


The issue – how to make complex cable’s cross section look like one simple wire electrically, and every wire sound the same?

Matching multiple wires into a complex structure isn’t easy to do well. The ideal cable is one wire that is exactly the same as the opposite polarity wire. To meet other objectives, we usually have several wires.

More small wires will make a nice big capacitor (wires with a dielectric between them) and trash reactive signal conversion to resistance products. Inductance will inversely follow capacitance, messing up the current delivery, and to get BOTH intrinsically LOW, you can’t go “whole hog” on the opposite variable. The two variables are tied together inversely. Rats! A suitable compromise must be reached? Yes, audio is a compromise, as we are seeing. BOTH L and C need to be low in value and good design manages this. Trade-offs for better sound can, and should, be logically explainable.

The less understood variable is Inductance. This variable is a big contributor to more wires. We all think, “capacitance” for audio. Realize that if we had NO inductance, we could separate the wires as much as we wanted and eventually have no capacitance (outer space actually has capacitance, so that’s impossible, too). The reduction of the magnetic fields by proper cable geometry reduces the inductance, allowing a larger wire center-to-center distance for low capacitance.  The LOWER the electromagnetic field, the LARGER the loop area can be (lower capacitance) for a given inductance and vice versa. Too many cables ignore getting the electromagnetic field as low as possible. The higher the current (speaker cables) the more field energy you need to eliminate.  Wires with low electromagnetic fields and small loop area have the lowest inductance.  ICONOCLAST’s speaker cable design balances out the wires’ proximity to one another so as to not “rob Peter to pay Paul”. The unique weave pattern increases the average wire C-C (Center to Center) distance creating a wire pattern that CANCELS the electromagnetic field while increasing the average spacing for low capacitance. Cancelling the field energy allows me to also lower inductance which would be impossible to do with JUST wire spacing for capacitance alone. No magic need apply.

EVERY wire in a cable has to be the same wire if you use superposition of the electromagnetic fields traveling down each wire. This is why symmetrical cable designs are used to efficiently remove reactive time alignment issues. I measured the reactive time based issues on other designs and they all came up short. Capacitance and Inductance have to be the same on EVERY wire to as tight a manufacturing standard as is possible. Multiple, and differing wire sizes are too complex to align things nearly as well. The signal’s SPEED has to be best matched at all frequencies and not just the physical wire length. The wire’s “signal length” is the problem. Use too many non-symmetrical, differing sized wires and this is all but near impossible to do with all the variables involved. I call this type of mixed wire cable, “cable in a cable”. The effect is a kindergarten lunchroom in the dark; a mess.

In passive cable, you can’t force the highs to go in the small wire and the lows in a bigger wire, and adjust the wire lengths to offset the VP changes. The ENTIRE spectrum goes into EVERY wire, so now we compound the time based issues. Only active electronics can separate the spectrum, and that’s a problem too.

Does ICONOCLAST remove the “cables in a cable” problem? Only one way to find out and that is to MEASURE them. The data is showing each polarity with 12 two wire BONDED pairs, 24 wires in each polarity, and 48 total wires in each cable;

175.3815 pF (X) +175.3815 pF = 180.1803 pF

175.3815 pF (X) + 171.4507 pF = 171.4507 pF

X = +2.74 % and – 2.29% variation between wires, or, they are ~ 97.5% the same.

I would say yes, I got it right.

For ICONOCLAST speaker cable I set my design goal at no more than 50 pF on capacitance and 0.1 uH/foot an inductance (45 pF and 0.08 uH typical). On the interconnect I set the goal at 12.5 pF and 0.16 uH (12.0 pF and 0.15 uH typical).  This is WITH connectivity and tested to prove it.

The complex electromagnetic designs of the RCA, XLR and speaker cables allow ICONOCLAST to exist. The RCA is the most pure electromagnetic equation that I have to work with and defines the interconnect cable problem. How do we reach the greatness that a PROPERLY designed RCA does in the XLR design (matched impedance / phase)? How can I convert the small signal world of the RCA and XLR into the large current world in the speaker cable (low inductance with still low capacitance)?


            The issue – is it mostly LOG linear so we can’t hear it?

If it is true that we can’t hear LINEAR attenuation (measured Rs values say there is non-linearity) or TIME based issues in audio cables, WHAT are we hearing with optimized designs, i.e. those that try to get L and C to near ZERO as we can and with low time based issues? The design goal difference in ICONOCLAST is TIME based and I’m not so sure that the inaudibility of difference values of 5-10 micro seconds is correct. Linear attenuation, I agree, is MUCH harder for the ear to pick out in typical cable lengths. I said LINEAR LOG type decay.

Rest assured, if there is snake oil in these products it sure looks like physics to me. All the above data is measured and real. The question remains, WHY do the cables SOUND so much better if TIME based issues aren’t audible? WHAT are we hearing, then? The reactive TIME altering L and C along with the VP change with respect to frequency seem to be the difference in cables, and audibly so. Linear attenuation can’t account for the differences. Series resistance says that that factor isn’t as linear as we’d like, either. There is a measurable difference in cables resistance across the audio band.

Is attenuation linear? I measured the Rs (series resistance), with respect to frequency, of ICONOCLAST and saw a significant CHANGE in attenuation with high quality R, L and C. Look at standard 1313A speaker, 10 AWG Zip cord style cable (red trace). ICONOCLAST flattens resistive non-linearity artifact, and the interconnects are both flat to 20 KHz human hearing test point.  Still, look at the UNITS; it isn’t a wall of lost energy above 20 KHz.



12.0 Low Pass Filter Effect

The issue – Cable is a low pass filter, and rolls off the frequency at the frequency of the filter’s cut-off; Fc.  How does this change what we hear?

I saved this one for dead last since it was even overlooked on my categorization of issues with audio cables. You’ll see why in a moment.

Here is the basic circuit. There is actually a SMALL inductor in series with the resistor but notice that it doesn’t appear in the equation that defines how the filter will behave, and is omitted. There are circuits that involve larger inductors, and unless we have a resistor to ground, they won’t apply to “cable” filters. Well, decent cable anyway.

The capacitor is a reactive device, like I’ve mentioned before, so its properties change with frequency as does an inductor. A capacitor eventually looks like a short to ground (capacitive reactance value keeps changing) at higher frequencies so the signal energy takes the path of least resistance through the capacitor to ground. R is in Ohms when capacitance is in picofarads (pF).

The good thing about almost ALL audio cables is that the roll-off properties of the filter are WAY above the audio band. Yes, a first order filter will change the PHASE at the -3dB attenuation point by 45 degrees, and time based distortions are more audible than the roll-off attenuation. First order filter attenuation nor phase changes are going to be an issue, theoretically.

Typical ICONOCLAST™ R, L and C Variables

Capacitance12.5 pF/ft12.5 pF/ft45 pF/ft
Inductance.15 μH/ft.15 μH/ft 0.08 μH/ft
Resistance32Ω/Mft 14Ω/Mft 1.15Ω/Mft

The RCA shield “goes away” as it is such a low resistance in series with the center wire, leaving essentially the center wire DCR.

The XLR uses TWO 25 AWG wires in parallel for each polarity, so the resistance is HALF the two wires, or about the same as a ~22 AWG wire.  

 Calculating Fc we arrive at;

15.5 GHz for the 5 foot RCA.

36.4 GHz for the 5 foot XLR.

15 GHz for the 10 foot Speaker Cable.

The real problem with cable is that it can load down the output op-amps with too high capacitance and change the frequency response and possibly phase response. Some really high capacitance or high inductance speaker cables can bug the heck out of power amplifier output stages, too. But these problems aren’t filter problems, but bulk capacitive or inductive loading problems on the output circuits.

All circuits “push back” below their operating region into the pass band but a rule of thumb is to keep the fc pass band 10X or more above the circuit’s operating frequency. We surely are meeting that requirement with any decent cable, even zip cord.


Many outside this sub-discipline of engineering will STILL insist that electromagnetic field time management and time alignment are not important, and that only the bulk R, L and C matter. The ear is a time domain instrument and readily time aligns the signal to the natural world we live in. EVERY effort was made to pay attention to TIME domain issues in audio cables and attenuation non-linear artifact. There are a myriad of ways to lose track of TIME, and an audio cable is not a good place to make mistakes.

Consider all the measured and factual information above on cable design and then ask yourself why cables sound different. Why wouldn’t they sound different given how complex  it all is? True, poorly made cables all fall into a bunch of warm and soft sounding products. Elevate the engineering and they indeed measure different. The above is 100% true for ALL cables, if I may add. If I mischaracterized a topic then, of course, only my cables are affected! All the cable designs in the ICONOCLAST line are under US patents.

I hope my cables bring years of enjoyment to you, and NEVER a feeling of complacency in what was provided to enhance your hobby’s (mine too!) pleasure. The search is constant to try to align TIME based issues to arrive at the best sound possible. The bad layers of the onion can’t be removed, but the order and thicknesses can be altered. Signal coherence is both arrival time and amplitude time dependent. Passive cable won’t allow perfection, just a lot of hard work to manage the ill effects that Mother Nature threw our way.

“Sound Design Creates Sound Performance”, and this means driving down all measurable variables to the lowest possible balance we can achieve. Does this make better sounding cable?


Galen Gareis
Principal Product Engineer
ICONOCLAST Design Engineer

When lending an ear, there’s no reason to leave one’s brain behind

Do not all charms fly
At the mere touch of cold philosophy?
There was an awful rainbow once in heaven:
We know her woof, her texture; she is given
In the dull catalogue of common things.
Philosophy will clip an Angel’s wings,
Conquer all mysteries by rule and line,
Empty the haunted air, and gnomed mine—
Unweave a rainbow, as it erewhile made
The tender-person’d Lamia melt into a shade

-Keats, from Lamia

Sound, as we know it, is subjective experience.  It comes at us through the space around us and we feel its force even as all manner of other spectral phenomena pass through us without being felt.  This subjective experience is, to us, everything — it is everything that music ever is or could be. 

But audio reproduction, regardless of how we experience it, is a heavily-engineered phenomenon.  That this is so leads to a kind of split-mindedness in audio discussion.  So often we see the subjective discussed only in terms of the subjective, and the objectively measurable discussed only in terms of objective measurement.  That electrical engineers have not, as a class, been receptive to some phenomena as reported by users has led, for some in the audiophile world, to a kind of obsession with subjectivity itself — as though all of those lovely sound waves could be delivered to us without the “cold philosophy” of electrical engineering that makes audio reproduction possible.    

This needn’t be so.  The problems of any aspect of audio engineering must be solved in order to render audio in the first place, so we cannot complain, like Keats, that the knowledge of how it is done will unweave the rainbow.  Indeed, sound reproduction is a bit like unweaving and then reweaving that rainbow, and there’s no getting around the objective issues involved in optimizing that subjective experience. 

A great deal of audiophile wire and cable design has been done by enthusiasts of one sort or another, listening to various designs and trying to figure out what they can.  Iconoclast Cable is fundamentally different: Galen Gareis, our cable designer, was a distinguished wire and cable engineer with Belden long before Iconoclast was even a notion, with decades of experience developing cables for a wide variety of applications.  When he embarked on the process of designing the Iconoclast speaker and interconnect cables, he had the use of the Belden Engineering Center, a facility with exceptional capabilities and equipment. 

Yes, the process of design is meant to optimize the subjective listening experience.  But how does one know what to optimize?  When are four conductors, or forty-eight, better than one, and why?  How?  And are these things practical to manufacture?  Galen, with his grasp of wire and cable theory, was as well situated as anyone has ever been to figure these things out. 

In this blog, and on this site, we endeavor to take you behind the curtain to understand our product — to understand WHY Iconoclast cables are made the way they are and what Galen did to bring them about.  Instead of “pay no attention to that man behind the curtain,” we’re committed to openly discussing and showing our work.  A product like this should have a sound technical rationale, and in addition to Galen’s papers on this site, we will bring you fresh posts from Galen’s work from time to time to talk about issues in wire and cable generally, and Iconoclast specifically. 

But, back to that rainbow we’ve been unweaving and reweaving.  While the realm of engineering is where the rubber meets the road in terms of delivering the sound, there’s no substitute for listening.  That is, after all, the point, and ultimately the only point.

We invite you to have a listen to Iconoclast.  But we won’t ask you to leave your brain at the door when you do.  We also invite you to read Galen’s papers on the design of Iconoclast, and if you have questions, let us know.