Shields and Grounding

BACKGROUND: There is always discussion on how to ground a shield. The answer lies in what the worst-case noise situation is. It isn’t always the same answer. Do you leave one end open or ground both ends? If you ground only one ends, which end?

We can look at how the interference behaves to answer these questions and they follow understood electrical properties within swept frequency regions.

BODY: The first step is to understand what a shield is doing, and how. In the simplest terms, a shield creates two separate “electrical” environments, one on each side of the shield. One side is measured as a RATIO of the field’s intensity relative to the other side, in dB.

The shields we are working with are ONLY effective with an EM wave that is predominantly “E” field in nature versus “B” field, or magnetic. Magnetic fields are not shielded or blocked with conductive shields, but need a shield material that blocks magnetic flux lines or a low permeability material…think “material a magnet will stick to.” Those kinds of material allow magnetic flux lines an easier path than through air. We can capture and re-route the flux lines in low permeability materials. For this discussion, we will look at mostly electric field shields, stuff that conducts electricity.

What is an ideal electrical shield? It is a shield that 100% blocks electrical energy at the surface of the shield, and that has infinitely low resistance. Shields don’t have infinitely low resistance and they block electrical energy at differing impedance based on the skin depth of the shield at a specific frequency.

We can measure the effectiveness of a shield across frequency with a transfer impedance plot. This is measured in milli-ohm/meter. It describes the resistance we can expect a shield to have, and thus the ratio of the fields energy in the shield based on the CURRENT the resistance causes to flow at that frequency…and it is not linear.

The graph below shows the frequency-dependent nature of a shield. A perfect shield would have NO RESISTANCE and both ends would be identical and thus seem like a SINGLE point of reference to a flow of current. Since we have zero resistance across the shield, we can’t have current flow caused by the shields. We CAN have current flow between the two points connected at the ends of the shield. In a “perfect” world the GROUND at both ends is the same potential and thus, an ideal shield has ZERO current flow, and is a measure of the POTENTIAL on one side of the shield relative to the other, in dB.

Since we don’t have a perfect world, Transfer Impedance describes what to expect at frequencies based on the shield’s impedance, and how that shield resistance creates a CURRENT flow and thus a voltage (shield resistance times the shield’s impedance = a voltage). When we have different resistances at each end of a shield we have current flow.

Shield Type5 MHz10MHz50MHz100MHz500MHz
Bonded Foil +60% braid2015112050
Quad Shield 60% +40% Braid
Tri-Shield+80% Braid10.60.10.22
Bonded Foil +95% Braid10.50.080.091

The chart below is what JUST the shield impedance looks like for a set length of cable at lower frequencies. We see the same non-linear behavior of shield and frequencies.

A SEED (Shield Effectiveness Evaluation Device per IEC 61196-1) test shows the dB relationship to a Lower Shield Impedance. Series number 5 with a 95% coverage 45-degree braid and Duofoil tape is clearly superior.

Okay, we can see a shield is not perfect, and not linear. So how does this say what to do with each end of a shield? We have to weigh the CHOICE of HOW the shield WORKS to decide our fate.

–     If you have ideal grounds and meet IEEE bus bar grounding (see the 568C.2 or later grounding specifications) limits it means BOTH ends can be grounded and the shield current will inductively couple less interference than the shield ATTENUATES through its material composition.

–     If we have severe ground differential, we can induce a strong current in the shield that CAN, if the shield’s resistance is higher, induce noise into the core that is WORSE than if we disconnect one end. We convert our shield to an antenna, not a shield!

–     An antenna does NOT create two separate environments between them with the ratio of one measured to the other. One end of the antenna is infinite impedance (the open end) with the other end at ground. The antenna “wire” is as close to zero impedance as possible in order to NOT attenuate the antenna’s signal going to ground. The signal won’t go to the open end, but seeks the lowest potential in the circuit.

We trade the noise caused by a poor ground resistance potential between shield ends for the induced noise in an antenna’s wire parallel to the signal wires that induces a voltage based on the antenna resistance. In an antenna type ground, it is best to ground the SEND end, as the SIGNAL on the internal wires is as LARGE as possible relative to the antenna signal, improving the signal to noise between the two.

An often-ignored aspect of shields is HOW to ground one at lower frequency versus RF. There is a big difference and again, it is based on the shield characteristics at each frequency.

The charts below are derived from TWO slightly differing MODELS of RF shield inductive reactance resistance. I have this paper for those interested. But, the data is the same message in that as frequencies increase, the shield reactance goes UP. This necessitates a FULL 360 ground at the shield termination point in RF circuits. This is why good RF connectors are fully capturing the shield all the way around the cable. On your RF digital cables, use 360 degree grounds for the best true shielding.

SUMMARY – Most systems will have proper GROUND differentials between them and thus have near ZERO shield current. The shield relative to the signal wires will ATTENUATE outside interference. When you have poor grounds, it may be beneficial to unhook the “receiver” end of the shield and hope that the induced antenna current voltage is less severe than the induced voltage caused by differential shield ground potentials. This should be a SECOND choice, not the first. A properly working shield, by design, has a KNOWN shield dB rating that can be trusted in a proper electrical circuit.

An antenna ground’s induced voltage onto the cable is not fully described and is dependent on the GROUND proximity point and shield’s distance from the signal wires. In severe situations, it may be the best choice to mitigate noise to the lowest possible reference value as it is pretty hard to REMOVE a shield already on a cable. Some, such as coaxial cables, can’t be mitigated and need to be properly designed to EXCEED the ground differential by several orders of magnitude so as not to aggravate any ground differential.

ICONOCLAST will use double ground interconnect shields and proper DCR RCA grounds. Power cables should also use grounds at BOTH ends if you have a proper GROUND plane resistance such that ZERO current flows and thus you have ZERO induced voltage from differential current. An antenna type ground CREATES a differential in each end of the “antenna” by design (one end is ideally infinity the other is ideally zero) and is thus a second choice if you have known ground issues.

One last note, those heavy 10 AWG power or more cables, may provide benefit as they induce less ground differential resistance than smaller power cords as the ground wires are larger. The circuit may not need the power delivery of a larger cable, but the lower ground resistance values may be of benefit on longer runs in marginal power grid situations. This will improve a shield’s current to nearer zero across frequencies. The dB isolation numbers values are for a proper shielded system with IEEE and TIA compliant shield differentials.

The Transfer Impedance numbers are between two-reference point probes on a shield, and DO NOT need the ground potential differences for characterizations.

Iconoclast Gen2 interconnect update

NOTE: This paper was originally written prior to the introduction of Iconoclast Gen2 interconnects, so while some references are to the future, that future is now….

BACKGROUND:  To possibly improve the performance of the XLR, to maybe achieve even lower L and C,  we would need to revise the current design…and it will jump up the electromagnetic complexity. The balanced of L and C would shift some but the coherence will improve substantially.

Changing the “conductor” to a four insulated wire structure will lower INDUCTANCE through signal phase cancellation. The star quad arrangement will retain CMRR for NOISE reduction. Four smaller wires will improve PHASE, and lower wire loop DCR to mitigate ground loops.

                                                                                                PROTOTYPE IMPROVED DESIGN:

Capacitance is the DISTANCE between the plates (wires) and dielectric material(s).

Inductance is two-fold;

  • The electromagnetic field cancellation.
  • The loop area between the wires changes inductance.
  • For inductance dielectric doesn’t really matter, inductance is DISTANCE.

We will have the same nearly loop area in the design (C-C distance is the same) but each conductor in the new design will further remove signal electromagnetic fields based on the cancellation geometry. Inductance should drop compared to the single wire conductor system.

The capacitance requires the same meticulous attention paid to as the group dielectric. Since we are keeping the cable the same size so the capacitance HAS TO go up as we have more wires parallel to a dielectric (the center X-filler, beading and outer tube) , and closer to the dielectrics. The conductor size and dielectric determine the final size. The added wires and X-filler close to the wires are the main contributors to the required capacitance increase. But, lower inductance improves PHASE shift, and your ear is most sensitive to.

The current coherence, the main objective of the design with minimal L and C changes, is based on the skin depth penetration changes going from 1 x 0.018” wire to 4 x 0.010” wire for each conductor.

4 wire “conductor”

Technically, four -wire per conductor will increase capacitance some as we have more wires parallel to a dielectric, but the  current coherence improves substantially, time aligning the low to high frequencies.

18214.4 μ inches = 18.2 mils @ one skin depth.

One skin depth is defined as when the surface current is 37% smaller going into the wire.  If we had a wire that was 18.2 mils in size, the CENTER of the wire would have only 37% of the current measured on its surface.

Skin depth equation (below) is a squared equation, so removing wire depth rapidly increases the inner current magnitude. Dropping from 20 mils to 10 mils is a 4X improvement in current coherence.

The very good, and easier to make, current design does NOT use electromagnetic signal field reduction technology I developed for the speaker cables in the series 1 signal leads. The current XLR design relies on reduced loop area and uses AIR to reduce the capacitance to a minimum for a given tighter spacing to achieve inductance.  The better the dielectric the CLOSER I can physically locate the signal wires for a given capacitance, thus lowering Inductance. The size of the wires determines the current coherence, and with more uniform effect of the dielectric around each wire with respect to frequency. The smaller the wire, the more uniform the velocity of propagation from low to high frequencies.

An XLR cable’s external noise utilizes CMRR based on all four noise signals being equal on each wire and which cancels those noise signals in a star quad design through electromagnetic field cancellation. If we look at the four wires, and using the right hand rule (current out of the page). All the external noise currents in the wire go CCW around each wire suspended in space. All the electromagnetic fields cancel adjacent to any wire and across from any other wire. All the fields superimposed onto one another forming a nearly ideal cancellation circuit. Nearly perfect because stray magnetic fields would extends OUTSIDE the four wires and reinforces the field. A first approximation says that this doesn’t happen. The stronger fields are closets to the wire and cancel most aggressively. Theoretical outer fields are weak, and don’t reinforce nearly as much as the inner fields cancel.



We DO NOT see this nearly “perfect” rejection of signal magnetic fields to reduce the inductance in the signal fields for series 1 RCA or XLR cable. We have a PLUS and MINUS balanced signal current direction whose fields are only partially cancelled. The partial field partial cancellation RAISES the inductance above “zero” theoretically as we have a stronger field, and separated by the distance needed to lower capacitance with any dielectric. The old design has ~36% higher inductance, and thus worse PHASE shift than series II (0.015 uh/foot is reduced to 0.11 uh/foot nominal). Lowering inductance directly lowers phase. See the QED phase analysis measurements on a variety of cable;

QED – The Sound of Science

OVERALL 4 x 1 wire XLR CMRR INTERNAL (signal energy)

The two MINUS fields cancel between themselves.

The two PLUS fields cancel between themselves.

But a MINUS to PLUS field REINFORCES the overall magnetic field.

The reinforcement makes the field stronger and the loop area effect worse.

BODY –To make improvements, we need to reduce the signal electromagnetic field to ZERO, in theory, both from an external interference view AND an internal electromagnetic conductor view. To do this, we need to BALANCE the music signal by SPLITTING each of the four SIGNAL wires into FOUR, or sixteen separate wires.

Making this critical change will theoretically remove the signal field currents that interact with the loop, creating inductance. It will also significantly improve the dielectric group and Phase delay by forcing the dielectric to be seen more uniformly across the 20-20KHz frequency range with smaller wires.

To keep capacitance low for a given loop area, we need to use AIR around the wires, and to make sure any plastics that touch the wire are super low dielectric constant materials (FEP mini X-filler and external FEP bead wire).  This is why the wires have to be BARE copper with NO insulation around them.  Only the tangential surface of the FEP filler and FEP beading, the rest is air. Capacitance is dielectric AND distance related where Inductance is distance and electromagnetic field strength.

Each of the four wire will be shorted together to make the typical four wires in a star quad. The wires are 10-mil diameter 30 AWG for a total CMA of 4 x 102 = 400 CMA. I used 4 x 0.018” in iconoclast for a total 18 x 18 = 324 CMA for each signal wire. 400 CMA is slightly lower DCR than the current design improving attenuation and mitigated ground loop voltages.

The collateral filler is foam FEP to manage capacitance. The power carrying signal braid should also be as far away as possible from the internal signal wire QUAD structure to lower the ground plane inside the cable, lowering capacitance. This means making the outer belting thickness under the braid to the best fit for an XLR connector, but not too big as the reduction in capacitance is a squared law variable, once a threshold is reached, more is not too beneficial.

The X-filler is FEP, as would the 30-mil beading wrapped around the QUAD wire to lower the dielectric nearest the wire where it is most critical. The material issues all control capacitance, not inductance.

The overall belt is solid FEP, with a 36 AWG BC braid and a drain wire. A final solid FEP jacket finished the cable.

CORE                    0.230”
BELTING             0.030”
BRAID                  0.015”
JACKET                0.030”
TOTAL                  0.305”

Does it really work on initial Capacitance and Inductance measurements? The final design using the ICONOCLAST™ all FEP design for ultimate performance appraisals measured as follows:


Lab Rqst-177575
Sample ID – 60156Y (PDC2842)

Requestor – Galen Gareis
Report Generation Date – 22 June 2017

Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Meas:  18.23 pF/ft

Inductance @ 1 kHz per ELP 424, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Meas:  0.10 µH/ft


Lab Rqst – 177587
Sample ID – PDC2842

Requestor – Galen Gareis
Report Generation Date – 29 June 2017

Capacitance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Cap @ 1 kHz Spec:  10.5 pF/ft max

Meas:  17.48 pF/ft

Inductance @ 1 kHz per ELP 423, Agilent E4980 Precision LCR Meter, Belden 4TP Cap/Ind Test Fixture

Meas:  0.10 µH/ft

Velocity of Propagation (VOP) per ELP 392, HP8751A Network Analyzer, HP VEE Instrument Control Software with Velocity of Propagation program and a GPIB card installed.

Meas:  85.3%

4×4 Design1×4 Ref. Design
231 pF/12.67′ = 18.23pF/ft12.5 pF/ft
1.29 μH/12.67′ = 0.10 μH/ft0.15 μH/ft

To really get better XLR performance, both loop area and the field cancellation technology need to be leveraged, with the latter being most critical. The capacitance is all about materials and DISTANCE between them. Improving inductive field cancellation has the added, and significant, benefit of improving signal coherence through four smaller wires and phase with lower inductance while improving attenuation performance.

A cable with smaller signal wires and better coherence, low inductance (better phase) and slightly higher capacitance will sound better sounding than a cable with larger signal wire and less coherence, higher inductance and lower capacitance…as long as capacitance isn’t too high!

The prototype run does indeed lower inductance with the expected rise in capacitance.

0.15 uH/foot (X) + X = 0.1 uH/foot, X= 33% lower.
Or the current design is 50% higher.

12.5 pF/foot (X) + X = 18.23 pF/foot = 45.8% higher
Or the current design is 31.4% lower.

Since the original design is working from such low L and C numbers, the percentages are not really illustrating the advantages of the improved signal coherence with much smaller wires, and an advantage that should play out in audible performance.  The -3dB first order filter frequency is still well above the audio band so first order filter phase distortion is not going to be an issue. What must be the major contributor is coherence with the smaller wires. Rs response, while lower, is hard to quantify.

Rs (swept frequency resistance) Values

The 4×4 XLR lowers swept Rs (proximity effect) values significantly, and flattens the high-end linearity. Can you HEAR that improvement, over the single the wire design? The truth is BOTH are superimposed when the wire is used, and pushing the XLR designs to as near perfection is certainly a better and better design. The lower DCR is evident in the trace compared to the 1×4 25 AWG wire as is the flatter upper frequency measurements.

The RCA interconnect has also been updated with the new 1×4 (ONE wire made with FOUR conductors) design. The reactive variables will track with frequency like the single wire designs, but map to the altered L and C values.

The following table shows the effects of changing the wire size and number. The 4 x 4 has almost the same CMA as a single 22 AWG, but 1.82 times more total circumference, which shows up only at increased frequencies. The lowest frequencies are essentially DCR.

The maximum Rs is lower with the 4×4 design. Beta test feedback from customers on the 4×4 has been extremely positive and, consistent with the numbers, shows this revision to be a significant upgrade from the original Iconoclast design for analog applications.


The interconnect cables of  a given wire design (single to single and verses quad to quad) have essentially the same loop DCR values.

From the Rs chart above at DC, we see;

4×1 XLR and 1×1 RCA are 34.11 and 39.19 Milli-ohms/foot respectively.

4×4 XLR and 1×4 RCA are 27.53 and 28.79 Milli-ohms/foot respectively.

How was this done? The double braid on the RCA was necessary to mitigate ground loop DCR variation between sources, and the DCR was designed to be near a “free” return path for loop[ DCR. The loop resistance ios the braid plus the  conductor. But, the braid DCR is so low that the loop DCR is pretty much the RCA center conductor. This is true for eother design.

The XLR DOUBLES the number of conductors in each leg as a star quad. This reduces the DCR to one-half the conductor’s value. Thus the two pairs in parallel are the same DCR as a single conductor.

This was also done on purpose to make sure that the RCA’s loop performance was as good as the XLR, and that the RCA BRAID was essentially a ZERO DCR return path between grounds. If the RCA braid was insufficient DCR, we would see more divergence between the two singel ended and balanced design.


The measured XLR electricals are very good, and follow design theory perfectly. ICONOCLAST once again shows that proper engineering fundamentals are paramount to performance. Sound Design Creates Sound Performance!

XLR Design Brief

1.0 Conductors
1.1 Copper Size
2.0 Dielectric Materials
3.0 Dielectric geometry
4.0 Shield Material and design considerations
5.0 Jacket design and material considerations

1.0 Conductors.

  1. Copper Size.

BOTH of the copper conductor and size considerations were answered when we started the RCA cable. We don’t want to change the current coherence with a differing conductor diameter if we are to mirror the reactive variables, too. We need the same exact wire to shield reactive L and C parameters in each cable in the end configuration design. The geometry of each cable is entirely different so how to you do that? That is, assuming you want to match the RCA and XLR properties and maintain the same signal quality…and we certainly do.  There is no reason to copy a bad sounding RCA cable when designing an XLR, so the RCA is designed FIRST.

  • Dielectric material(s).

One difference in the XLR is that we are going to use FOUR wires in a star quad configuration. (Note: in our “Gen 2” XLR product, there are sixteen wires — four wires making a star quad in place of each single wire in the design shown below.  For more detail, see the last paper in this series.)  Four wire XLR cables use two cross-connected wires for each polarity, which doubles-up the wire AWG for lower attenuation. Two 25 AWG have the DCR of a single 22 AWG yet has way better signal coherence by using smaller wire.

I could have used a cheaper and easier two wire XLR design but the inductive and signal coherence benefits of a star quad are too good to pass-up. If I can get the materials and quad design to achieve a high enough level of performance it is a better cable design.

Star quads have a higher degree of CMRR (Common Mode Rejection Ratio) when properly signal balanced. There are three primary reasons for this;

  • The two or four wire stranding “twist”.
  • The differential encoding.
  • The outer shield properties, but only at RF frequencies.

Two wires of a star quad are a “positive” voltage, and two wires are a “negative” voltage (180 degrees out of phase), hence the term “balanced”.  If the cable were a teeter-totter, it would sit level. Some call this differential mode since each signal is equal but different.

Differential Mode Transmission
Perfect Wire Balance Equals Less Noise

In the example above we show two wires, but the system is the same in a star quad. The signal we WANT is encoded as +2 volts and -2 volts. The noise can’t “change its spots” relative the cable’s twisted pairs and shows up as the same voltage on each wire, +1V noise in this example. The TWIST ratio helps make sure that the wires see the noise the same amount of time and this is vital to the function of the circuit.

Here is where the balance is so important; the signal IDEALLY becomes the superposition of ALL the voltages, or +3 volts and -1 volt. No more, no less. The signal voltages are STILL exactly 4 volts “apart” from each other; +2 to -2 with no noise and +3 to -1 volts with the noise.  The signals are fed into a difference amplifier that, you guessed it, looks at the “difference” between the two voltages and see’s 4 volts with, or without, the noise. The noise is absent in a perfect world at the difference amplifier’s output.

In order to do this, every wire has to be presented to the noise in the exact same way via the cable twist and has to be the same length so the signal stays TIME aligned down the wire and has to have the same attenuation. The difference amplifiers need to be nulled perfectly between gain halves. Believe it or not, this gets done really well with good quality products.

The control tolerance of the copper is 0.0005”, so attenuation issues are mitigated and CUB (Capacitance UnBalance) tests insure we see MIL standard quality in the finished cable. All quality types of copper can be used in the XLR design. It is the overall structure that is the most “magic” and not as much the copper itself, although the copper draw process does influence the sound.

We have several variables that aren’t present in a coaxial cable design to contend with;

  • CUB, Capacitance Unbalance or, each wire shows a differing capacitance to ground.
  • DCR unbalance, each wire has to be the same DCR.
  • CMRR remainder, the differential signs have to NULL to the exact same point neither above nor below reference ground.
  1. Dielectric geometry.

Lots of words, time for a picture;


The above CAD drawing is what we have inside our XLR design so far (well, I ignored two wires in the drawing).

Remember I wanted to make L and C reactive variables EXACTLY the same for each cable with EXACTLY the same wire size and draw science? What else do we know? I also said that CAPACITANCE is sensitive to the distance to a conductive plate area, and that means ALL the way around the wire. The coaxial cable is easy; we purposefully put a ground around the wire at a known distance that defines the capacitance ground plane reference distance and inductive loop area.

In the coaxial cable, the center of the wire to the inside of the tube is 0.098” / 2 = .049”.  Ok, so what? This is what. The capacitance is a squared law property and predominantly sees the ground closest to the wire. The shield on the opposite side of the XLR cable, to a first approximation, falls a way. We actually measure the capacitance BETWEEN the two cross paired wires but the ground location still influences the capacitance. Also, we have four wires that are capacitors.

This doesn’t “sound” good, does it? We have four times as many wires and all have capacitance. Somehow this is supposed to come out around 12 pF/foot (with connectors), same as the RCA!

Now for the inductance part, L. Inductance is loop area defined. It could care less about the dielectric, but the graph above shows a HUGE ~0.170” loop area! How is THAT going to get to the 0.15 uH/foot inductance of the coaxial cable? I could make something up, but that isn’t as neat as what’s really going on.

To get capacitance as low as I need it to be to match the coaxial cables, I use DISTANCE between the wires. And yes, this DIRECTLY sets what the inductance will do…hold on a minute.  By using AIR, I can set the C-C of the wires to meet my capacitance target needed for the final tested value with two cross wires connected and tested between them. AIR lessens this distance for a given value of C so I can also manage inductance now. For inductance, L, the smaller the wire loop area the better for a given value of total capacitance. Air gets me far closer than any other dielectric.

How much air? Well, EXACTLY the same as the coaxial cables! How do we do that? The standard answer is, “very carefully”. Let’s look at a drawing;


Still, so what? Yep, I agree, until we compare this area to the area in in the RCA cable air dielectric; 0.00754 in2. OK it isn’t exact; I missed by ~0.000009” in2.  I use the exact same thread design around each identical wire so it’s all the same area in the chamber as in the RCA.

Let’s do some reality checking as to what it SHOULD be based on MEASUREMENTS and calculations.

  • We have the EXACT (can I say that as close as it is?) same velocity of propagation based on the composite (air and plastic inside the ground plane) dielectric; 87% at RF reference.
  • I measured the IMPEDANCE at RF @ 100 ohms, same as the coaxial cable.
  • The dielectric constant can be calculated and from that the VP, VP = 1/ SQRT (E).
  •  And from that composite dielectric I also know what the capacitance has to be.

Capacitance (remember that chart on dielectric value and capacitance earlier?) is directly linked to the group dielectric constant. I know VP, and I know the impedance, so I can calculate the capacitance and then get the dielectric constant from that.

101670 / (C * 87) = 100 ohm

C = 11.68 pF/foot.

What does the cable actually measure on capacitance? The chart below shows 11.767 pF/foot. Notice that the capacitance values between each of any two wires has to be ~ 5 pF/foot to “double-up” the two wires capacitance and still to arrive at a final ~11 pF/foot! Yep, that’s LOW capacitance. Capacitance adds in parallel so this is a significant issue when a design uses four wires.

Below is the measured and calculated imbalance of the capacitance between 1-3 and 2-4 cross wires’ conductors as a “pair”; 2.02% unbalance, very low.

We seem to have the capacitance and VP looking much like the coaxial cable. Remember, measurements include ALL the approximations in the soup.

So what about inductance with that WAY larger loop area? Isn’t that going to really kill this thing?  No, because of some properties of magnetic fields. Magnetic fields CANCEL if they see each other in OPPOSITE directions. Inductance is the “reactance” or “resistance” to instantaneous flow of current. If we can REDUCE the magnetic field lines, we can directly reduce the measured inductance.

We also know from the basic equations that DISTANCE between the two wires is important. Keeping BOTH distance and magnetic field line magnitude small lowers inductance, and removes the noise.

The picture below shows what’s going on…sort of. For now, we’ll pretend the field’s ONLY go “inwards”, or inside the wire, and stop there (they don’t). If the lines that extend outside each wire do the OPPOSITE as the field INSIDE the wires, they reinforce the field!  It is generally accepted that the flux lines concentrate substantially BETWEEN the wires.

If we draw arrows that represent the DIRECTION of the circumferential magnetic field waves AROUND each wire we get what is shown below for a NOISE signal hitting the wire. We have TWO different voltage polarities so we have TWO different current directions for the SIGNAL, but the NOISE is the SAME direction in all the wires.

If you grasp a wire with all four of your fingers, and point your right hand THUMB in the CURRENT direction, your fingers will point in the field’s circumferential direction around each wire. The arrows are a “part” of the noise current field lines “inside” the four-wire group.

NOISE FIELDS (all the same direction)

Where the arrows are OPPOSITE each other in direction between any two wires, the field lines cancel. For NOISE every field theoretically cancels. ADJACENT or ACROSS from any TWO wires we induce field cancellation with a star quad design.

For the SIGNAL, we now have TWO equal but opposite current directions.

This allows larger wire-to-wire spacing in order to lower capacitance and also keeps inductance low. Inductance is managed with field line cancellation geometry.

Now we know why I didn’t use a two-wire system, you can’t manage CMRR.

Let’s look at the situation for the signal. Below is a simple picture of the field cancellation between four wires with opposite polarities wired as a star quad.

SIGNAL FIELDS (opposite directions)

   Minus = Current INTO the page (CW rotation)

   Plus = Current OUT of the page (CCW rotation)          

Reduce Signal Loop Area to Reduce Inductance

What do we see? The all the signal field lines DO NOT cancel. Adjacent wires reinforce, and opposites wires cancel. Reducing loop area is the best way to manage inductance because we can’t cancel all of the field lines, only some of them. This theoretical field relationship limits the ability to reduce capacitance for a given inductance. Using low dielectric constant materials to lower capacitance (Air!) allows closer spacing needed for low inductance.

Is there a design that can, in theory do BOTH, reduce signal and noise fields to “zero”? Within the limits of DESIGN, yes there is. The ICONOCLAST series II reduces both noise and signal field cancellation. The wires, in practice aren’t EXCATLY the same distance apart and EXACTLY the same resistance, so we say “in theory”. But, reducing the nose to 1000 or more times less and reducing the inductance 27% is indeed achievable.

So, after all that explaining, how does the star quad ICONOCLAST cable measure up? Tests at 1 KHz show the following values below. The inductance between the two cross wire pairs of the star quad are 0.15 uH/foot inductance…same as the RCA.

So what does the “reactive” picture look like comparing the RCA and XLR? How close are they to being the same? This swept test is the real deal. There are no approximations to fudge.

What we see above is impedance / phase for the XLR and RCA superimposed one on top the other. Note that there are four separate lines. We have two identical cables with exceptionally low reactive variables.

  • Shield material and design considerations.

There is yet one last thing to consider in the XLR design; the outer shield. A 95% BC (Bare Copper) braid is used. Audio cables are not RF designs, and the braid shield will NOT shield low frequency magnetic interference. The CMRR of the XLR is going to do that for us. Excellent CUB, DCR unbalance and twist ratio all aid CMRR. The braid DOES knock down RFI by 80 dB, so that’s a given. The shield isolation @ RF mitigates NULL balance at high frequencies only.

Like it or not, 20-20K is a predominantly magnetic field frequency range where the B-fields decay at a ratio of 1/x^3. DISTANCE is the best solution for isolation of cables with magnetic properties.

5.0 Jacket design and material considerations.

All ICONOCLAST cables use FEP as the jacket to reduce UV sensitivity, plasticizer migration and provide chemical resistance.  The cables are designed to last decades.

I hope that this design summary of ICONOCLAST RCA and XLR interconnect cables shows how important good design is for ALL your audio cables, and that every manufacturer has to manage all the same variables to produce these results. There is little “magic” in the design of good cables. There are indeed tertiary variables that we can’t measure, but those should not influence the ones we can measure, or at least not excessively so. Mother Nature abhors complexity, so the better managed the known variables in a cable are, the more properly it may highlight “unknowns.”  To put it another way, the more we put knowns into their proper place, the better we may distinguish the effects of the unknown. Wire draw science, for instance, can be heard better, and more fairly, in a superior electromagnetic design.

Belden appreciates your interest in how quality interconnects are made, and how / why ICONOCLAST RCA and XLR cables were physically derived as you see them in their production form. We have no special sauce or magic in our products, and I think that the cables perform as well as they do BECAUSE we did not design around “unknowns” and then make it appear as though we had unique influence on those unknowns in the design.

Truly low R, L and C cables are difficult to make when consideration is given to all three variables to manage them in a truly balanced fashion. The designs can be frustratingly simple looking but hard to manufacture, as processes are pushed to the limits of current capabilities. Belden’s focus is to make real measured values as low, and properly balanced, as we can. ICONOCLAST interconnects represent the pinnacle of low frequency measurements and electrical balance between the RCA and XLR (same electromagnetic properties).

The next design analysis will look at the SPEAKER cable.